Generating a tail current for a differential transistor pair using a capacitive device to project a current flowing through a current source device onto a node having a different voltage than the current source device

ABSTRACT

A current tail circuit and method for a differential transistor pair affords the capability of sensing an input differential signal having a low common mode voltage when using, for example, an NMOS differential transistor pair. A current source device and a capacitor may be employed to provide at the common node of the differential transistor pair what appears to be a constant current source connected to a “negative voltage.” In one embodiment particularly useful when using an NMOS differential pair, one terminal of a capacitor is precharged to VDD and the other terminal is precharged to VSS (i.e., ground). When the amplifier needs to sense its differential input signal, a control signal turns off precharge transistors and couples the capacitor terminal previously precharged to VSS to the common-source node of a differential transistor pair. The capacitor terminal previously precharged to VDD is driven toward VSS, preferably with a controlled current source, which couples the common-source node of the differential pair from VSS toward a voltage below VSS. As soon as the common-source node voltage is low enough for at least one side of the differential pair to conduct a current substantially equal to the controlled current source, the common-source node voltage is substantially clamped at that voltage. The actual voltage resulting on the common-source node depends on the transistor characteristics, the particular voltages present on the gates of the differential pair, and the magnitude of the controlled current source. For some operating conditions, this voltage may be above ground rather than below ground, but the “tail” current of the differential pair nonetheless remains equal to the magnitude of current through the device driving the other terminal of the capacitor from VDD toward VSS.

CROSS-REFERENCE TO RELATED APPLICATION

This application claims the benefit of U.S. Provisional Application No.60/120,032, filed Feb. 13, 1999, which application is incorporatedherein by reference in its entirety.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to amplifier circuits, and particularly tothose amplifier circuits providing a tail current source coupled to thecommonly-connected node of a differential transistor pair.

2. Description of Related Art

As is well known in the art, a basic differential amplifierconfiguration is advantageously implemented using a pair of transistorsconfigured as a differential pair, and using a constant current sourcein the “tail” of the differential pair. For example, a pair of NMOStransistors form a differential transistor pair when their respectivesource terminals are connected in a common node, or common-source node,and their respective gate terminals are connected to respective inputterminals (e.g., for receiving a differential input signal). As anotherexample, a pair of bipolar transistors form a differential transistorpair when their respective emitter terminals are commonly-connected in acommon-emitter node, and their respective base terminals are connectedto respective input terminals. When using NMOS transistors in thedifferential transistor pair, a constant current source is frequentlyapproximated by a single N-channel transistor with a DC bias voltage onits gate having a value which is less than the nominal common modevoltage of the two input signals connected to the gates of thedifferential pair. Such a configuration assumes that the current tailtransistor remains saturated, which requires its drain voltage to exceedits gate voltage less a threshold. But in a traditional differentialpair configuration, the drain voltage of the current tail transistor(assuming it's the same node as the common-source node of thedifferential pair) must be lower in voltage than the higher of the twoinput signals less a threshold voltage for any current to flow througheither of the differential pair transistors.

As a result, it is exceedingly difficult to use a traditional NMOSdifferential pair, having threshold voltages optimized for operation ofthe remaining circuitry of the integrated circuit, to sense a signalhaving a very low common-mode voltage which approaches or is below thethreshold voltage (relative to, for example, the lower power supplyvoltage). To achieve such a low common mode input voltage, designers mayuse a PMOS differential pair biased toward an upper power supply. Such aPMOS differential transistor pair is usually slower than an NMOS pair,and also suffers from an analogous problem when presented with an inputsignal having a high common mode voltage which approaches (e.g., withina PMOS threshold voltage) the upper power supply voltage. As a result,designers frequently use both NMOS and PMOS differential transistorpairs, each connected to receive the differential input signal, with thecurrents through each pair summed to produce an output signal. Each ofthese techniques frequently adds complexity, degrades performance, orresults in unwanted amplifier characteristics.

SUMMARY OF THE INVENTION

To afford the capability of sensing an input differential signal havinga low common mode voltage when using, for example, an NMOS differentialtransistor pair, a current source device and a capacitor may be employedto provide at the common node of the differential transistor pair whatappears to be a constant current source connected to a “negativevoltage.”

In one embodiment particularly useful when using NMOS transistors in adifferential transistor pair, one terminal of a capacitor (e.g., an MOStransistor configured as a capacitor) is precharged to VDD and the otherterminal is precharged to VSS. When the amplifier needs to sense itsdifferential input signal, a control signal turns off prechargetransistors and couples the capacitor terminal previously at VSS to thecommon-source node of a differential transistor pair. The capacitorterminal previously precharged to VDD is driven toward VSS, preferablyusing a controlled current source, which couples the common-source nodeof the differential transistor pair from VSS toward a voltage below VSS.As soon as the common-source node voltage is low enough for at least oneside of the differential pair to conduct a current substantially equalto that of controlled current source, the common source node voltage issubstantially clamped at that voltage. The actual voltage resulting onthe common-source node depends on the transistor characteristics, theparticular voltages present on the gates of the differential pair, andthe magnitude of the controlled current source. For some operatingconditions, this voltage may be above ground rather than below ground,but the “tail” current of the differential pair nonetheless remainsequal to the magnitude of current through the device driving the otherend of the capacitor from VDD toward VSS.

No device configured to function as a constant current source is neededbetween the differential transistor pair and the capacitor because thetail current of the differential transistor pair is controlled by andsubstantially equal to the current discharging the other end of thecapacitor, which current flows through the capacitor as a displacementcurrent. If the discharging current is a constant current, the tailcurrent is also a constant current. In this example, discharging a nodepreviously precharged to VDD with a constant current toward VSS may beaccomplished using straightforward circuits, such as an NMOS switchtransistor in series with an NMOS current mirror transistor.

In a broader embodiment of the invention useful in an integrated circuitincluding a differential transistor pair having a common node connectinga respective first current handling terminal of each transistor withinthe pair, a method of providing a tail current for the differentialtransistor pair includes: (1) precharging a capacitive device having afirst terminal and a second terminal; (2) providing a current path fromthe common node of the differential transistor pair to the firstterminal of the capacitive device; and (3) driving, with a firstcurrent, the voltage of the second terminal of the precharged capacitivedevice in a direction to capacitively couple the first terminal and, byway of the current path, the common node to respective voltagessufficient in magnitude and polarity to cause a current, substantiallyequal to the first current, to flow collectively through one or bothtransistors of the differential transistor pair; (4) wherein the firstcurrent flows as a displacement current through the capacitive deviceand is projected onto the common node, thereby providing a dynamic tailcurrent for the differential transistor pair substantially equal inmagnitude to the first current.

For certain values of input voltages applied to the differentialtransistor pair, certain transistor characteristics of the differentialtransistor pair, and certain magnitudes of the first current, the firstterminal of the capacitive device may assume a voltage outside a rangeof voltages bounded by the highest power supply voltage and the lowestpower supply voltage operably received by the integrated circuit. Inother cases, the first terminal of the capacitive device may assume avoltage within the range of voltages bounded by the highest power supplyvoltage and the lowest power supply voltage. In certain embodiments, thecapacitive device is disposed in close local proximity to, and uniquelyassociated with, the differential transistor pair. In other embodiments,the capacitive device is disposed in close local proximity to, and alsouniquely associated with, two separate differential transistor pairs,and each differential transistor pair may include a respective gatingcircuit connecting the respective common node thereof to the firstterminal of the capacitive device. In certain embodiments, theconnection path consists of a direct wired connection between the commonnode and the first terminal of the capacitive device, or, for otherembodiments, may include at least one transistor connecting the commonnode to the first terminal of the capacitive device.

In an apparatus embodiment of the present invention, an integratedcircuit includes: (1) a first transistor and a second transistorconfigured as a differential transistor pair having a common nodeconnecting respective first current handling terminals of eachtransistor within the pair; (2) capacitive means having a first terminaland a second terminal; (3) means for precharging the capacitive means toa voltage thereacross; (4) means for providing a current path from thecommon node of the differential transistor pair to the first terminal ofthe capacitive means; and (5) means for driving, with a first current,the voltage of the second terminal of the precharged capacitive means ina direction to capacitively couple the first terminal and, by way of thecurrent path, the common node to respective voltages sufficient inmagnitude and polarity to cause a current, substantially equal to thefirst current, to flow collectively through one or both transistors ofthe differential transistor pair; (6) wherein the first current flows asa displacement current through the capacitive means and is projectedonto the common node, thereby providing a dynamic tail current for thedifferential transistor pair substantially equal in magnitude to thefirst current.

In another apparatus embodiment of the present invention, an integratedcircuit includes: (1) a first transistor and a second transistorconfigured as a differential transistor pair having a common nodeconnecting respective first current handling terminals of eachtransistor within the pair; (2) a capacitive device having a firstterminal and a second terminal; (3) a precharge circuit for precharging,during a first time period, the capacitive device to an initial voltagethereacross, said first terminal of the capacitive device being coupledto a source of a first voltage and said second terminal of thecapacitive device being coupled to a source of a second voltage; (4) acurrent path from the common node of the differential transistor pair tothe first terminal of the capacitive device; and (5) a driver circuitresponsive to a first control signal, for driving, with a first currentduring a second time period, the voltage of the second terminal of thecapacitive device in a direction to capacitively couple the firstterminal and, by way of the current path, the common node to respectivevoltages sufficient in magnitude and polarity to cause a current,substantially equal in magnitude to the first current, to flowcollectively through one or both transistors of the differentialtransistor pair; (6) wherein the first current flows as a displacementcurrent through the capacitive device and is projected onto the commonnode, thereby providing a dynamic tail current for the differentialtransistor pair substantially equal in magnitude to the first current.

The present invention may advantageously be used in amplifier circuitsincorporated into a variety of useful circuits, including a linearamplifier circuit, a latching amplifier circuit having a separate latchenable input, and a latching amplifier circuit having no separate latchenable input, where latching occurs as a result of the differentialoutput signal developed by the differential transistor pair.Advantageous embodiments may be fashioned using any of a variety oftransistor structures, including field effect transistors as well asbipolar junction transistors.

As an example, the present invention may be advantageously used in aread amplifier in the read path of a dynamic memory array to developsignal on a generic I/O line before bit line sensing has occurred. Theinputs for such a read amplifier may be connected to the bit lines, thesense amplifier nodes, a local I/O line serving, for example, a few bitline pairs, or a local output line similarly serving, for example, a fewbit line pairs. The present invention may also be advantageously used inan external input buffer to sense external input signals having a smallsignal swing and a common-mode voltage near the lower power supply. Oneinput for such an external input buffer may be coupled to receive anexternal input signal, with the other input coupled to receive areference voltage or a complementary external input signal. Such anexternal input buffer may advantageously be strobed or enabled by acontrol signal generated relative to an external clock signal receivedby the integrated circuit.

The scope of the present invention in its many embodiments is defined inthe appended claims. Nonetheless, the invention and its many featuresand advantages may be more fully appreciated in the context of exemplaryimplementations disclosed and described herein which combine one or moreembodiments of the invention with other concepts, architectures,circuits, and structures to achieve significantly higher performancethan previously achievable. For example, a high performance dynamicmemory array architecture is disclosed in several embodiments, alongwith various embodiments of associated supporting circuitry, whichafford performance approaching that usually associated with staticmemory arrays.

In an exemplary embodiment an 18 MBit memory array includes four banksof arrays, each including thirty-two array blocks. Each array blockincludes 128 horizontally-arranged row lines (i.e., word lines) and 1152(1024×9/8) vertically-arranged columns. Most internal circuitry operatesusing a single positive power supply voltage, VDD, and the referencevoltage VSS (i.e., “ground”). Each column is implemented as acomplementary folded bit line pair. Four independent row decoders areprovided respectively for the four banks, and are physically arranged intwo pairs, thus forming two splines, one spline located between the leftpair of memory banks, and the other spline located between the rightpair of memory banks. Latching input buffers for address and controlinputs are located within each of the splines and are connected torespective input pads by horizontally arranged input wires runningthrough the memory banks. Two input buffers are provided for each inputpad, one located in each spline. Clock lines used to strobe the variousinputs are arranged vertically, running through each spline. An R-Ccompensation circuit between each input wire and the correspondinglatching input buffer located in the particular spline nearest itsrespective input pad provides a delay to the “upstream” buffer whichcompensates for the additional wiring delay in reaching the “downstream”buffer, and which allows all of the latching input buffers to be drivenby phase-aligned clock signals, and still achieve a very narrow worstcase setup and hold time over all such inputs. The use of a separateinput buffer in each spline for each address and control input,requiring additional interconnect wire to connect each input pad to itsinput buffer in the “far” spline (above and beyond the interconnect wireto connect each input pad to its input buffer in the “near” spline),increases the input capacitance of each address and control input to thechip (which input capacitance, of course, must be driven by the sourceof the external signal). However, the complementary internal outputs foreach such input buffer may be buffered immediately by self-resettingbuffers, and need only drive decoder and/or control circuitry locallywithin the same spline. Thus, the total capacitive loading on thecomplementary outputs of each buffer are advantageously reduced and aremore balanced between the various buffers.

The row decoder uses predecoding to reduce the total line capacitancedriven during an active cycle. The final stages of the row decoderincludes an N-channel tree configuration driven by VDD-level (i.e.,VSS-to-VDD level) pre-decoded address signals to select and discharge toVSS a particular decode node which was precharged to VPP. Subsequentbuffering stages provide a final 1-of-4 decode and drive the selectedword line to a VPP voltage that is substantially independent of VDD,rather than driving the selected word line to VDD or to a voltage whichis a ratio of VDD. There are no race conditions within the decoder, eventhough it accomplishes a level shifting from VDD-level signals toVPP-level word lines.

The VPP voltage is internally generated by a charge pump type circuitand its output is a substantially fixed voltage independent of processand environmental corner which is regulated with respect to VSS (i.e.,ground). For typical operating voltage, the VPP voltage is somewhathigher than VDD, although at low operating voltage the VPP voltage maybe substantially higher than VDD, while at high operating voltage, theVPP voltage may be similar in magnitude to the VDD voltage. Preferablythe VPP voltage is chosen to be near the maximum voltage that the fieldeffect transistors (FETs) can safely tolerate. Since the VPP isregulated to be substantially independent of variations in the VDDvoltage, the VPP level is advantageously at a higher voltage than wouldotherwise be safe, and tolerances in the VPP voltage level which wouldotherwise be necessary to account for variations in the VDD level areunnecessary.

If the semiconductor technology allows, transistors which are exposed tothe VPP level (e.g., transistors whose gate terminal is driven at anytime to the VPP level while the source or drain terminal might be atground, such as the memory array access transistors and various arrayselect transistors, or those transistors whose drain or source terminalis driven at any time to the VPP level while the gate terminal might beat ground) are preferably implemented using a thicker gate dielectricthan the majority of the other transistors which are never exposed tosuch a high differential voltage across gate-to-drain or gate-to-sourceterminals. Moreover, it is also preferable to limit the voltage acrossany transistor using the thin gate dielectric to no more than VDD.Transistors exposed to any voltage which is greater than the VDD levelare preferably implemented with the thick gate dielectric and arelimited in voltage to the VPP level, which is a fixed voltagesubstantially independent of the VDD voltage. Consequently, transistorsexposed to such internally “boosted” voltages need only withstand arelatively fixed, predictable voltage level (e.g., by using a bandgapreference in the circuit which regulates the VPP voltage) and do notneed to withstand even higher voltages which might otherwise be producedby a “boosted” voltage generator whose output voltage is a ratio of VDD(e.g., 1.5×VDD). The voltage across the memory cell capacitors islimited to less than one-half VDD (e.g., limited to about 1.0 volts forcertain embodiments). A third dielectric material, thinner than the“thin” capacitor dielectric required for typical DRAM memory cells(which must normally support a voltage of one-half the maximum allowedVDD voltage) may be advantageously used to fabricate the memory cellcapacitors to provide additional storage capacitance per unit area.

Within each memory bank, a row of sense amplifiers is implemented in theholes between each pair of array blocks. Each sense amplifier is sharedbetween two pairs of bit lines—one pair located within the array blockabove the sense amplifier and the other pair located within the arrayblock below the sense amplifier. The complementary internal nodes withineach sense amplifier are respectively connected to the true andcomplement bit lines above the sense amplifier by a first pair ofN-channel array select transistors whose gates are driven to VSS (toisolate the sense amplifier nodes from the bit line pair) or driven toVPP (to connect the sense amplifier nodes to the bit line pair), and arefurther connected to the pair of bit lines below the sense amplifier bya second pair of array select transistors whose gates are likewiseswitchable from VSS to VPP. A row of sense amplifiers is implementedabove the top array block and another row of sense amplifiers isimplemented below the bottom array block of the given memory bank, whichserve half of the bit lines within the top and bottom array blocks,respectively. For any particular array block, half of the bit line pairsare served by a sense amplifier located above the array block, and theremaining half are served by a sense amplifier located below the arrayblock. A pair of array select transistors having a gate voltageswitchable between VSS and VPP connects any given pair of bit lines tothe complementary internal sense amplifier nodes within thecorresponding sense amplifier.

An amplifier in the read path is used to develop signal on a generic I/Oline before bit line sensing has occurred. Such a generic I/O line mayinclude a global output line, a column line, or an I/O line. Thisamplifier may be connected to the bit lines, the sense amplifier nodes,a local I/O line serving, for example, a few bit line pairs, or a localoutput line similarly serving, for example, a few bit line pairs. If theread amplifier inputs are connected directly to the bit line senseamplifier nodes (i.e., one read amplifier per bit line sense amplifier),the column select function may be advantageously used to enable theamplifier for the selected column, while if the read amplifier inputsare connected to local output or I/O lines (i.e., one read amplifier pergroup of bit line sense amplifiers), the column select function may beused to couple the selected bit line sense amplifier to the local outputor I/O lines. If the common mode voltage of the read amplifier inputnodes is so low that current flow through the tail of an N-channeldifferential pair cannot be assured for all voltage or process corners,the amplifier may incorporate a coupling circuit to capacitively couplethe tail of the differential pair downward, preferably using acontrolled current source, to approximate a constant current source to anegative supply voltage.

In a certain embodiment, each read amplifier's inputs are connected tothe internal nodes of a corresponding bit line sense amplifier. Therespective outputs of a group of read amplifiers are connected in commonto a horizontally-arranged differential pair of local output lines. Onesuch amplifier is enabled at a time by column select circuitry todevelop signal on the pair of local output lines. A second stageamplifier then further buffers this signal and drives a pair ofvertically-arranged global output lines. The global output lines extendthe full height of the memory bank, with half preferably extendingbeyond the memory bank to I/O circuits above the memory bank, with theremaining half extending beyond the memory bank to I/O circuits belowthe memory bank. In certain embodiments, the second stage amplifier mayalso include a multiplexer to choose between two different pairs oflocal output lines (e.g., a first pair of local output lines serving 8sense amplifiers located to the left of the second stage amplifier, anda second pair of local output lines serving 8 sense amplifiers locatedto the right of the second stage amplifier).

The word lines within the array blocks may be implemented in apolysilicon layer and strapped using a later-processed metal layer toreduce word line delays. Such word line straps are preferablyimplemented using two different layers of metal (preferably the two“lowest” layers, metal-1 and metal-2) in order to match the word linepitch without requiring any distributed buffers or final decode buffers.The read amplifiers used to sense a local output line and subsequentlydrive a global output line may be advantageously located above word linestraps where a break in the memory cell stepping already occurs. Thisallows the read amplifier block to more readily be laid out in thecenter of a group of bit line sense amplifier and column selectcircuits. As such, the bit line sense amplifier pitch may be slightlyless than twice the column pitch (recalling that half of the bit linesense amplifiers are above the array block and the remaining half belowthe array block).

The bit line sense amplifiers each are implemented using a full CMOScross-coupled latch. To sense the signal on a pair of bit lines, boththe cross-coupled N-channel pair of transistors (i.e., the NMOS senseamplifier) and the cross-coupled P-channel pair of transistors (i.e.,the PMOS sense amplifier) which form the CMOS sense amplifier areenabled at substantially the same time. The NMOS sense amplifier drivesthe bit line having a lower voltage toward VSS, while the PMOS senseamplifier drives the bit line having a higher voltage toward VDD. Ifenabled a sufficiently long time, the lower bit line substantiallyreaches VSS and the higher bit line would be driven substantially allthe way to VDD. However, the PMOS sensing is terminated before thehigher bit line substantially reaches the full VDD voltage. This allowsthe bit line to quickly be driven to a high level without having to waitfor the “exponential tail” if it were driven all the way to VDD. Theinternal sense amplifier nodes and the near end of the bit lines areactually driven above and overshoot the final high bit line “restore”level (e.g., 2.0 volts for a device operating at a VDD of 2.5 volts)before the PMOS sensing is terminated, whereas the far end of the highbit lines have not yet reached the final high bit line “restore” levelwhen the PMOS sensing is terminated. Then, after the PMOS sensing isterminated, charge is shared between the near end and far end of the bitlines, thus speeding up the far end reaching the final high bit line“restore” level because the effective time constant of the resistive bitline is cut in half.

Since the word line and array select lines are left high for some timeeven after the PMOS sense amplifier is turned off, charge sharingbetween the sense amplifier nodes, the near and far ends of the bitlines, and the memory cell storage node itself contribute to determiningthe final high restore level which is “written” back into the selectedmemory cell. When compared to having a full VDD level on a high bitline, the relatively low final “high” bit line voltage (e.g., 2.0 volts)transfers into the selected memory cell more quickly due to the highergate-to-source voltage of the memory cell access transistor.

The NMOS sensing is preferably continued, even after the PMOS sensinghas stopped, to more adequately drive the bit line having the lowervoltage (the “low-going” bit line) to a substantially full VSS level.This ensures that, if the selected memory cell happens to be coupled tothe low-going bit line, a substantially full VSS level is restored intothe selected memory cell. This also ensures that all the low-going bitlines (not just those having a selected memory cell connected thereto)are filly discharged before, at the end of the cycle, the high and lowbit lines share their charge to set the bit line equilibrate voltage.The selected word line (which is driven when active to the VPP level) isthen brought low as the NMOS sensing is terminated, after which thearray block is automatically taken into precharge.

Timing circuitry is used to time the simultaneous start of both NMOS andPMOS sensing relative to the timing of the selected word line beingdriven high, to time the end of PMOS sensing, and to time thesimultaneous end of NMOS sensing and the selected word line beingbrought low. The PMOS sense timing duration may be designed to decreaseas the VDD voltage increases to ensure a written high level which issubstantially independent of VDD, even over process and temperaturecorners. For example, the timing may be set to ensure a written highlevel on the high bit line (and into the selected memory cell) of about2.0 volts for a device having a VDD voltage range from 2.3 to 2.9 volts.Such a PMOS sense timing generator may be accomplished by using a dummybit line and sense amplifier structure (activated substantially beforethe main sense amplifiers are activated), detecting when the PMOSsensing needs to be turned off to achieve a final high voltage of about2.0 volts on the dummy sense amplifier and bit line structure; thenbuffering this timing signal to control the turn off time of the PMOSsense enable signals for the regular sense amplifiers within the memoryarrays. The PMOS timing may alternatively be accomplished using a stringof inverters powered at a voltage a fixed amount below VDD, or by othertechniques to achieve a timing which is a combination of severalvariables, such as power supply voltage VDD, bandgap voltage, transistorthreshold voltage and transconductance, temperature, or others.

In a preferred embodiment, the sense amplifier timing circuitry producesthree main timing signals. The first timing signal is used to control,relative to the timing of the selected word line being driven high, thesimultaneous start of both the NMOS and PMOS sensing. A second timingsignal is used to control, relative to the simultaneous start of NMOSand PMOS sensing, the duration of the PMOS sensing, and a third timingsignal is used to control, relative to the end of the PMOS sensing, whento simultaneously end the NMOS sensing and bring the selected word lineback low. Each of these timing signals is independently generated,although the circuitry used for each may share portions with another.These three timing signals define three timing intervals. The timinginterval “t₁” begins with the selected word line being driven high andends with the simultaneously start of both the NMOS and PMOS sensing(i.e., the timing interval “t₁” is the amount of time the selected wordline is high before sensing). The timing interval “t₂” extends from thesimultaneous start of NMOS and PMOS sensing to the end of PMOS sensing(i.e., the timing interval “t₂” is the duration of the PMOS sensing).The timing interval “t₃” extends from the end of the PMOS sensing to thesimultaneous end of the NMOS sensing and discharge of the selected wordline (i.e., the timing interval “t₃” is the amount of time the word lineremains high after the end of PMOS sensing).

The timing interval t₁ essentially controls how much signal from thememory cell reaches the sense amplifier before starting the NMOS andPMOS sensing. A short t₁ may not provide enough time for all the chargein a selected memory cell to fully share with the charge on the bit lineand sense amplifier nodes, and consequently the sense amplifier beginsto sense with less signal than would be developed if, alternatively, alonger t₁ were configured. A longer t₁ increases operating margins atthe expense of increased cycle time. Similarly, the timing interval t₂essentially controls how much charge is driven onto the high-going senseamplifier node, bit line, and memory cell during sensing. Increasing t₂increases the voltage stored into the memory cell, but also increasesthe bit line equilibrate voltage when charge is later shared betweentrue and complement bit lines (and sense amplifier nodes). A short t₂may not provide enough charge to develop the desired restored high level(e.g., 2.0 volts) on the bit line and into a selected memory cell.Conversely, an excessively long t₂ timing may not increase the storedhigh level in the memory cell as much as it increases the bit lineequilibrate voltage, and thus may decrease the high level signalavailable for sensing, particularly at high VDD. The timing interval t₃essentially controls how much charge is shared between the senseamplifier node, the near end and far end of a high-going bit line (whichtypically is moderately resistive), and the memory cell. The resistanceof the NMOS memory cell access transistor is much higher when restoringa high level (due to its lower gate-to-source voltage) than whenrestoring a low level. The t₃ timing is constrained by the time neededto write a high voltage into the selected memory cell through theresistive bit line and further through the relatively high-resistancememory cell access transistor. A short t₃ may result in a worst casememory cell (one located at the “far” end of a bit line, furthest fromits bit line sense amplifier) being written to a restored high levelwhich is too low, for a given amount of “Q” transferred into the senseamplifiers (i.e., for the bit line equilibration voltage which resultsfrom the given amount of “Q”).

These timing intervals t₁, t₂, and t₃ may be collectively optimized on achip-by-chip basis. In a preferred embodiment, there may be sixteendifferent timing settings, each specifying a particular combination ofthe t₁, t₂, and t₃ timing intervals, ranging from very aggressive forhighest performance, to very relaxed for highest yield. For example, thetiming setting “1” may provide for the most aggressive (i.e., shortest)t₁ timing interval, the most aggressive (i.e., shortest) t₂ timinginterval, and the most aggressive (i.e., shortest) t₃ timing interval.The timing setting “16” may provide for the most relaxed t₁ timinginterval, the most relaxed t₂ timing interval, and the most relaxed t₃timing interval. Each incremental timing setting between “1” and “16” ispreferably optimized to incrementally increase, by a similar amount, thesignal available at the bit line sense amplifier just before sensing. Toaccomplish this, the timing setting “2” may increase the t₁ interval by200 ps compared to the “most aggressive” t₁ value of timing setting “1,”while keeping t₂ and t₃ unchanged (a 200 ps increase may be easilyachieved by adding two inverters to the logic path setting the timeinterval). The timing setting “3” may increase t₃ by 200 ps whilekeeping the same value of the t₁ and t₂ intervals as in timing setting“1.” Each successive low-numbered timing setting preferably increasesthe value of one of the three timing intervals t₁, t₂, and t₃ relativeto their values in the previous timing setting, while keeping theremaining two timing intervals unchanged. Higher numbered timingsettings may increase a given timing interval by increasingly largeramounts to maintain a similar increase in the signal available at thebit line sense amplifier just before sensing, or may increase more thanone of the three timing intervals. For example, the timing setting “15”may increase t₁ and t₃ each by 400 ps relative to the respectiveintervals in timing setting “14” (compared to a 200 ps increase in onlyt₃ between timing setting “2” and “3”).

The timing setting “8” is preferably optimized to provide a “nominal”value for each of the three timing intervals t₁, t₂, and t₃ which isexpected to be an appropriate setting for a typical device havingtypical transistor characteristics, typical sense amplifier offsetvoltage, typical bit line resistance, etc. Note that these “nominal”values of the timing intervals t₁, t₂, and t₃ are a function of theprocess corner. Higher bit line resistance, higher access transistorthreshold voltage, or lower VPP, for example, raise the nominal value ofeach of the t₁, t₂, and t₃ timing intervals which are called for bytiming setting “8.” For the preferred embodiment, the various timingsettings provide a variety of t₁ intervals, some shorter than nominaland others longer than nominal, and provide a variety of t₃ intervals,both shorter and longer than nominal. But since the duration of the PMOSsensing is so short for the nominal case, for some embodiments theshortest t₂ interval provided is the “nominal” value, and more relaxedt₂ intervals are provided for in the timing settings numbered above “8.”

During manufacture, this timing setting “8” is configured as the defaultsetting. During a special test mode (for example, at wafer sort) thetiming setting may be temporarily made more or less aggressive todetermine the window of operation for each chip. Some of the memorydevices are found to function correctly with very aggressive timing,while others require more relaxed timing. Then, during the fuse blowingsequence for redundancy, timing fuses may be also blown to permanentlymodify the default strobe timing. The timing setting is preferably setas aggressively as possible to enhance device performance, whilemaintaining adequate sense amplifier signal margins for reliability. Forexample, if a timing setting of “4” is the most aggressive timing forwhich a given device functions without error, then the device may beadvantageously fuse programmed to a timing setting of “6” to ensure someadditional operating margin (the signal to the bit line sense amplifiersincreasing as the timing setting increases). At a later test, such as atfinal test of a packaged device, the test mode may still be entered, andthe timing setting advanced from its then fuse programmed setting to amore aggressive setting, in order to further verify adequate senseamplifier margins on a chip-by-chip basis, independent of which actualtiming setting was fuse programmed into the device.

A two-dimensional grid of power buses is preferably implemented withineach memory bank, with large VDD and VSS buses arranged parallel to thebit lines and implemented in a higher layer of metal (e.g., the toplayer), vertically passing above the bit lines. Filter capacitors arelocated at the ends of each array block as well as at the top and bottomof each memory bank to help provide additional bypass capacitance towithstand the large current spikes which occur during sensing. Thesefilter capacitors, as well as other filter capacitors implementedelsewhere within the device, are preferably implemented using multiple,independent capacitors which are individually de-coupled andautomatically switched out of the circuit if, at any time, more than apredetermined leakage current is detected automatically by the memorydevice as flowing through a given capacitor (i.e., a “shorted”capacitor). The large metal buses allow this stored charge to reach thetwo selected rows of sense amplifiers (i.e., located in the holes aboveand below the selected array block) with very little voltage drop, andallow the sense amplifiers to latch quickly and provide a good VSS lowlevel.

The bit lines are equilibrated together to achieve an equilibrationvoltage on the bit lines, for a preferred embodiment, of approximately1.0 volts. The bit lines are preferably equilibrated at both ends toreduce the required equilibrate time. The bit line equilibration voltageis coupled from all bit line pairs to a common node which may be sampledjust after equilibration and buffered (using a sample-and-holdamplifier) to drive the memory cell plate. Since the bit lineequilibration voltage is approximately one-half the written high level,the bit line equilibration voltage may also be sampled, compared to areference voltage (for example, a 1.0 volt reference), and any voltagedifference used to adjust the PMOS timing (and thereby adjust the finalwritten high level).

As stated above, the exemplary memory array is automatically taken backinto precharge without waiting for a control signal. In other words, oneedge of a clock causes the memory array to execute a useful cycle, thento automatically reset itself in preparation for a new cycle. Thisprecharge timing is relative to the beginning of the active cycle. Ofsignificance, this limits the amount of potential sub-threshold leakagethrough memory cell access transistors by limiting the time that any bitlines are at VSS. The precharging/equilibration is accomplished by usingtwo sets of signals—one is an automatically timed pulse, while the otherstays on until the start of the next cycle. For example, the bit linesense amplifiers are preferably equilibrated using two differentequilibrate signals. Both turn on automatically at the same time afterNMOS sensing is complete and the selected word line is brought low. Oneequilibrate signal is turned off by a timed pulse just when the bit lineequilibration is substantially complete (i.e., at the end of the activecycle), while the other equilibrate signal is turned off by the start ofthe subsequent cycle. The pulsed equilibrate signal drives much largerinternal capacitive loads, such as large equilibration devices, whilethe non-pulsed equilibrate signal drives fewer and/or much smallerdevices which indeed assist the larger pulsed equilibrate devices inequilibrating the various nodes. However, the smaller devices arelargely included as “keepers” to maintain the equilibration until thenext active cycle. As such, the total capacitance of the variousequilibration signal lines which must be discharged (i.e., brought low)at the start of new cycle is greatly reduced and can be accomplishedwith less delay after the initiating control signal, and the performanceis enhanced. For relaxed clock cycle times, the pulsed equilibratesignal falls automatically at the end of a cycle, while the non-pulsedequilibrate signal stays high until the next cycle selecting this arrayblock is initiated. However, for a clock cycle time which approaches thefastest possible cycle time for a given device, the non-pulsedequilibrate signal for the newly selected array block may be dischargedby the initiation of the next cycle at substantially the same time asthe pulsed equilibrate signal for the previously selected array block isdischarged automatically at the end of the previous cycle. To savepower, the non-pulsed equilibrate signal for only the selected arrayblock and supporting circuitry is brought to VSS at the start of anactive cycle, and all others remain inactive at VDD throughout theactive cycle. Similarly, the pulsed equilibrate signal for only theselected array block and supporting circuitry is actually pulsed at theend of an active cycle, while all others remain inactive at VSS.

During an internal write operation, the exemplary device contains writecircuitry that supplies a small differential voltage to the senseamplifier before bit line sensing, the polarity of the voltage dependingon the data to be written. The circuitry furthermore “swallows” thevoltage otherwise developed in the sense amplifier by the selectedmemory cell. Then, during their normal latching, the bit line senseamplifiers then “write” the level into the memory cell. Because of aninternal write queue, the data to be written is already available whenthe actual internal write operation is started. In preparation for thecurrent write operation, this data is preferably driven onto the globalinput lines late in the previous write operation, and then coupled tothe selected sense amplifier by column select circuitry fairly early inthe current write operation, before latching the bit line senseamplifiers. The magnitude of the write signal coupled onto the senseamplifier nodes is kept small to reduce power consumption and to reducedisturbance to the neighboring bit lines and sense amplifiers which arenot being written. Preferably, the magnitude of the write signalimparted onto any given sense amplifier node is no higher than thatnormally developed during a read operation, so that coupling to theneighboring bit lines and sense amplifiers is no worse than during aread operation. The global input lines serving the next word to bewritten are equilibrated after each write operation, preferably to thebit line equilibration voltage, and driven to the new data state for thenext write operation, even if the next write operation is not the nextcycle. Moreover, the differential voltage on the global input linesserving the next word to be written is equilibrated away (in a writecycle) after bit line sensing has started and the column select linesare inactive (i.e., during the later stages of bit line sensing), andthen driven to reflect the new write data for the following write cyclebefore the bit lines have finished equilibrating, rather than drivingthese data input signals during the early part of bit line sensing whensuch movement could disturb the bit line sensing. The global input linesthen dynamically float until needed by the next write operation. Tohandle the possibility that the next write operation may be many cycleslater, the global input lines may be refreshed periodically (e.g., every256 external clock cycles, before any leakage current can substantiallymodify their voltage) by re-equilibrating and re-driving to ensure theproper magnitude of the write data signal for as long as necessary untilthe next write operation occurs.

By writing a dynamic memory array by “fooling” the sense amplifier andletting it actually restore the voltage levels onto the bit lines inaccordance with the data to be written, rather than in accordance withthe data previously in the selected memory cell, a write cycle takes thesame very short time as a read cycle, rather than the longer time thatwould be required by first sensing old data, then modifying it. Inaddition, a significant amount of power is saved by not having toover-power many sense amplifiers after they have already been latched.

During power-up, all the memory cells are initialized to a low voltageunder automatic internal control. Provision is made to allow every wordline to simultaneously go high, to force the node to which the bit linesare equilibrated to VSS, and to ensure that the bit line equilibrationand array select transistors are on. Since each sense amplifier is thencoupled to a common node at VSS by precharge signals, each bit line(both true and complement) is driven to VSS and all memory cells arelikewise forced to VSS, even if the word lines are no higher than athreshold voltage above VSS. At about the same time, the memory cellplate is established at a voltage near the eventual bit lineequilibration voltage (preferably around 1.0 volts) by other power-upcircuits, being careful to limit the current flow, which charges thecell plate, to an amount less than the output current of the substratebias charge pump (to prevent the substrate from coupling positively andcausing massive latchup from the diffused regions of each memory cell'sinternal node). Then, when normal cycles begin, the very first operationin the memory array occurs with memory array nodes (bit lines, cellplate) properly established, and all memory cells initialized at one ofthe two valid states (in this example, at VSS). The first cycles do nothave to try to sense memory cells having an initialized voltage near thebit line equilibration voltage, as would likely occur without such apower-up sequence due to coupling from the memory cell plate to thememory cells themselves as the memory cell plate reaches its normallevel at the bit line equilibration voltage of, for example, 1.0 volts.This prevents any bit line sense amplifiers which are not being writtenfrom spending time in a meta-stable state which, if allowed to occur,would affect the high level restored into the memory cells beingwritten, as well as the equilibrate voltage resulting on the bit lines.

During a read operation, signal developed on the bit lines by theselected memory cell is immediately buffered by the local output lineamplifier(s) before bit line sensing starts, and immediately starts todevelop signal on the pair of global output lines. For certainembodiments, the differential signal propagates through lines anddifferential amplifiers to the output buffers, whose first stage is alatching amplifier which is then strobed to detect, amplify, and latchthis signal. The timing of the strobe signal for this latching amplifier(which may be known as “t₄”) may be optimized on a chip-by-chip basis.There may be, for example, eight possible strobe timings, from veryaggressive to very relaxed. The device may be initially configured withan intermediate default strobe timing (e.g., having a value of “4,”where “1” is the most aggressive and “8” is the most relaxed), andduring a special test mode (for example, at wafer sort) the strobetiming may be made more or less aggressive to determine the window ofoperation for each chip. Then, during the fuse blowing sequence forredundancy, timing fuses may be also blown to modify the default strobetiming. The timing is modified to be as aggressive as possible whilemaintaining adequate margins for reliability. For example, if in thetest mode a t₄ timing of “2” is the fastest timing for which a givendevice functions without error, then the device may be advantageouslyfuse programmed to a t₄ timing of “3” or not altered to remain at “4” toensure sufficient operating margin. At a later test, such as at finaltest of a packaged device, the test mode may again be entered, and thet₄ timing advanced from its then fuse programmed setting to a moreaggressive setting (e.g., 1 or 2 settings faster than its new programmedtiming setting without needing to know the new programmed timingsetting), in order to further verify adequate operating margins on achip-by-chip basis, independent of which actual timing setting was fuseprogrammed into the device.

In an alternative embodiment of a memory array having a cycle time whichis long compared to its read access time, a latching global output lineamplifier may be strobed (at what was time t₄ in the earlier embodiment)to detect and amplify the signal on the pair of global output lines, andcommunicate the sensed data onward through output multiplexer circuitryand ultimately (if the particular global output line is selected) tooutput buffer circuitry. The timing of the global output line amplifiermay be selected to support both a flow-through configuration as well asa pipelined configuration. To support a fast flow-through access timespecification, the latching global output amplifier is aggressivelystrobed as soon as a predetermined amount of signal has developed on theglobal output lines. In this way, the data propagates to and isavailable at the outputs as quickly as possible. But with thisaggressive timing, some devices may fail. Conversely, when in thepipelined mode of operation, the global output latch timing is relaxedto more closely coincide with the global output signal peak, and thesensed data is provided to the output buffers for driving to the outputpins during the next cycle (using a PLL or delay-locked loop). Byaffording additional time for even more signal to develop on the globaloutput lines, a particular device which may be marginal or may even failat the fast t₄ timing of the flow-through mode may prove to haveadequate margin at the more relaxed timing of the pipelined mode, andmay be sold for use and guaranteed to operate only in the pipelined modeof operation.

Bit line crossover structures are advantageously used to achieve lowerworst case coupling, during both read or write operations, onto aparticular bit line pair from neighboring bit lines on either side.Because photolithographic guard cells are used at the edges of eacharrayed group of memory cells, there is a layout area penalty inproviding crossover structures including the required guard cells oneither side of each crossover structure. To reduce this area penalty, anovel crossover arrangement is employed, for certain embodiments, whichprovides a significant degree of noise (i.e., coupling) reduction whilerequiring only one crossover. Within each array block, eachcomplementary pair of bit lines runs vertically from the top to thebottom of the array block. The true bit line and complement bit line ofa first pair run adjacent to each other from the top to the bottom ofthe array block without any crossovers. The true bit line and complementbit line of a second pair do not run adjacent to each other, but insteadstraddle the first pair (i.e., both true and complement bit lines of thefirst pair lie between the true and complement bit lines of the secondpair), with a single crossover half-way down the second bit line pair(vertically in the middle of the array block). This crossoverarrangement repeats horizontally throughout each array block in groupsof two pairs of bit lines (four physical bit line wires). By using thiscrossover arrangement, only four groups of guard cells are required ineach array block—one each at the top and bottom of the array block, andone each at the top and bottom of the single crossover structure locatedin the vertical center of the array block.

The address and data for a write cycle are queued to eliminate deadcycles on the system data bus. In the exemplary embodiment operated inthe pipelined mode, the address for a read cycle is strobed during onecycle, and the corresponding data read from the selected memory cells isdriven onto the external data pins during a subsequent cycle. If anexternal write cycle follows immediately after an external read cycle,the write address may be presented to the address bus and strobed intothe memory device just like for a read cycle, but the externalbidirectional data bus is occupied with driving the data outcorresponding to an earlier external read cycle (by a number of cyclesdepending on the pipeline latency for a particular embodiment) andcannot be used to present the corresponding write data. Instead, thedata for the external write cycle is driven onto the data bus andpresented to the device during the cycle in which output data would haveappeared had the cycle been an external read cycle instead of anexternal write cycle. In this way, the address bus and the data bus areused every cycle, with no wasted cycles for either bus. Both the writeaddress and data are queued, the actual write operation to physicallystore the write data into the selected memory cells is postponed until asubsequent write cycle, which then, when executed, retires thepreviously received address and data from the write queue into thememory array. Read bypass circuitry is provided which allows datacorresponding to the address of the read cycle to be correctly read fromthe write queue whenever an earlier queued write directed to that sameaddress has not yet been retired.

In the exemplary embodiment, the internal data path is twice as wide(i.e., a “double word”) as the external I/O word width (i.e., the leastsignificant address bit selects one of the two possible 36-bit words),and a significant degree of internal power consumption is saved bymerging external write cycles when sequential write addresses occur. Theaddress of a given external write cycle is stored and compared to theaddress of the next external write cycle. If the selected memory cellsto be written in both external write cycles correspond to the samephysical word line and the same column within the same array block ofthe same memory bank (i.e., differ in only the least significant addressbit), the internal write operation which would otherwise follow from thefirst external write cycle is delayed, and the data to be written isqueued and merged with the data to be written in the second externalwrite cycle. The write queue then “retires” both queued write requestsby performing a single internal write operation, simultaneously writingboth data words received in the first and second external write cycles.If the internal data path were wider than 72-bits, then more than two36-bit write cycles could be merged into a single internal writeoperation. For example, if the internal data path were 144-bits wide,then four 36-bit write cycles could conceivably be merged into a singleinternal write operation.

The exemplary embodiment includes a burst mode of operation whichprovides, during subsequent cycles, read or write access to sequentialaddressed memory cells relative to a received (i.e., “load”) address,without requiring such sequential addresses be presented to the device.Using the 72-bit wide (double word) organization of each memory bank,two 36-bit words are retrieved from the memory array in the first cycle.The second word is saved to present to the data outputs after the firstword is output. Because the exemplary device is organized into separatememory banks, a burst of four sequential words may transcend the addressboundaries between memory banks. Consequently, the exemplary deviceincludes provision for automatically initiating a load cycle in anothermemory bank during a burst cycle.

In certain embodiments, a dynamic memory array using the architectureand supporting circuits described above achieves random access cycles(each requiring a new random row access) at a sustained rate in excessof 200 MHz operation, even when each new row access is within the samearray block of the same memory bank.

The present invention may be better understood, and its numerousobjects, features, and advantages made even more apparent to thoseskilled in the art by referencing the detailed description andaccompanying drawings of the embodiments described below.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a memory device incorporating a dynamicmemory array which provides an exemplary embodiment for describing thefeatures and advantages of the present invention.

FIG. 2 is a block diagram of a portion of the memory array shown in FIG.1 and showing a plurality of array blocks having bit line pairsalternately connected to shared sense amplifiers above and below thearray block.

FIG. 3 is a simplified electrical schematic diagram conceptuallyillustrating the local and global I/O line aspects of the memory arrayshown in FIG. 1.

FIG. 4 is a detailed block diagram of an array block and the supportingcircuitry located above and below the array block, and particularly theconnection to the global I/O lines.

FIG. 5 is a block diagram of the memory array, and particularlyillustrating the alternating connections of the global I/O lines to thedata path multiplexer circuits located at the top and bottom of thememory array.

FIG. 6 is a block diagram illustrating the connection of every otherglobal I/O line to the data path multiplexer circuit located at thebottom of the memory array, and particularly illustrating the connectionmapping, for one embodiment, of each global I/O line to the respectivebit within the global data bus.

FIG. 7 is a schematic diagram of an embodiment of the sense amplifiercircuitry and arrangement shown previously in FIG. 4.

FIG. 8 is a schematic diagram of an embodiment of the local I/O toglobal I/O interface block shown previously in FIG. 4.

FIG. 9 is a schematic diagram of an embodiment of the global I/O toglobal data bus interface block shown previously in FIG. 6.

FIG. 10 is a schematic diagram of another embodiment of the supportingcircuitry located above and below the array block, and particularlyillustrating the use of a separate pair of local output lines betweenthe sense amplifiers and the global output lines, and the placement of aportion of the first stage amplifier within each of the senseamplifiers.

FIG. 11 is a schematic diagram of the embodiment shown in FIG. 10,illustrating the routing of the global input lines to each senseamplifier, and particularly illustrating the additional write circuitrywithin each of the sense amplifiers.

FIG. 12 is a schematic diagram of an address pre-decoding circuit usefulfor both row and column addresses.

FIG. 13 is a schematic diagram of an address circuit which receivesglobal pre-decoded row address lines and generates local pre-decoded rowaddress lines which correspond to and are local to a single array blockwithin a memory bank.

FIG. 14 is a schematic diagram of a row address decoder which receivesVDD-level pre-decoded local address signals, level shifts up to a VPPlevel typically above VDD, and buffers a word line from VSS to VPP, allwithout any race conditions.

FIG. 15 is a schematic diagram of another embodiment of a row addressdecoder portion for driving four word lines which is conceptuallysimilar to that shown in FIG. 14, but which includes a one-to-fourdecoder in the final word line buffers and two redundancy flip-flopsuseful for replacing pairs of word lines.

FIG. 16 is a schematic diagram of a support circuit for the row decodershown in FIG. 15 which during power-up disables all row lines and drivesthe common bit line equilibrate node to VSS.

FIG. 17A is a schematic diagram of a circuit which generates the upperand lower pulsed equilibrate signals and the upper and lower arrayselect signals for a row of sense amplifiers within a given hole betweenarray blocks.

FIG. 17B is a schematic diagram of a level-shifting inverter circuituseful for the circuitry shown in FIG. 17A.

FIG. 17C is a schematic diagram of a level-shifting AND-gate circuituseful for the circuitry shown in FIG. 17A.

FIG. 17D is a schematic diagram of another embodiment of alevel-shifting AND-gate circuit useful for the circuitry shown in FIG.17A.

FIG. 18 is a block diagram of a preferred embodiment of the VPPgenerator which produces a substantially fixed voltage, usually aboveVDD for most process corners, which is referenced to VSS.

FIG. 19 is a flow chart diagram of a preferred embodiment of thepower-up sequence for initializing all memory cells to a known datastate.

FIG. 20 is a block diagram of an embodiment having a memory array withfour distinct memory banks, having a first row decoder block between thefirst and second memory banks, and having a second row decoder blockbetween the third and fourth memory banks.

FIG. 21 is a block diagram of a memory bank within the embodiment shownin FIG. 20, illustrating an arrangement of global I/O lines providing afull 36-bit word while activating only one hole between array blocks,and two 36-bit words if two adjacent holes are activated, and which alsoallows reducing the length of the worst case global data bus.

FIG. 22 is a block diagram of the memory array within the embodimentshown in FIG. 20 and using the arrangement of global I/O lines as shownin FIG. 21, particularly illustrating a worst case global data bushaving a reduced length.

FIG. 22B is a schematic block diagram of an embodiment of the data pathbetween the global I/O lines and the external data output pin whichincorporates differential global data bus lines traversing horizontallyacross the top and bottom of the memory device.

FIG. 23 is a block diagram of a portion of a memory bank in accordancewith another embodiment of the invention which arranges the global I/Olines to provide contiguous data bytes.

FIG. 24 is a block diagram of a portion of a memory bank as illustratedin FIG. 21, illustrating the operation of various array block signalsfor the selected array block and the adjacent, non-selected arrayblocks.

FIG. 25 is a waveform diagram illustrating the major array and senseamplifier signals when reading a high from the selected memory cellwhile operating at a VDD of 2.9 volts.

FIG. 26 is a waveform diagram illustrating the major array and senseamplifier signals when reading a high from the selected memory cellwhile operating at a VDD of 2.3 volts.

FIG. 27 is a waveform diagram illustrating major read path data signals,corresponding to the other waveforms shown in FIG. 26, when reading ahigh from the selected memory cell while operating at a VDD of 2.3volts.

FIG. 28 is a waveform diagram illustrating additional major read pathwaveforms (some of which are shown in FIG. 27) but at a larger verticalscale to more readily perceive certain small amplitude signals.

FIG. 29 is a waveform diagram illustrating the major array and senseamplifier signals when reading a low from the selected memory cell whileoperating at a VDD of 2.3 volts.

FIG. 30 is a waveform diagram illustrating the major array and senseamplifier signals when writing a high into the selected memory cell(having a previously stored low) while operating at a VDD of 2.3 volts.

FIG. 31 is a waveform diagram illustrating the major array and senseamplifier signals when writing a low into the selected memory cell(having a previously stored high) while operating at a VDD of 2.3 volts.

FIG. 32 is a schematic diagram illustrating use of dual input buffersfor each address input for the memory array embodiment shown in FIG. 20,with one input buffer preferably located within the left spline, and theother input buffer located within the right spline, and furtherillustrates a timing compensation network for the internal clock signalwhich strobes each buffer, so that setup and hold times for both leftand right buffers are closely matched.

FIG. 33 is a block diagram of an embodiment of a feedback controlledcircuit for generating an internal clock signal which is phase andfrequency locked to an external clock signal, which is useful forstrobing other address and control input signals into the memory devicewith a setup and hold time window very closely aligned to the risingedge of the external clock.

FIG. 34 is a layout diagram of a preferred embodiment of atwo-dimensional power supply bus grid within a memory bank, andparticularly shows large VDD and VSS buses arranged vertically to runparallel to, and in a metal layer located above, the bit lines, andcovering substantially all the array block except for periodic groupingsof global input and output lines which may be implemented in the samelayer of metal as the VSS and VDD buses.

FIG. 35, labeled prior art, is a layout diagram of a well known bit linecross-over arrangement for reducing noise coupling from adjacent bitlines.

FIG. 36 is a layout diagram of a portion of a memory array block whichillustrates a preferred embodiment of an improved cross-over arrangementfor reducing noise coupling from adjacent bit lines within the arrayblock, which affords similar benefits as that shown in FIG. 33 whilereducing the area consumed by photolithographic guard cells.

FIG. 37 is a timing diagram of several external memory cyclesinterspersing external reads and writes for a representative embodimentof a memory device in accordance with the present invention, whichillustrates the latency of a read cycle, and the analogous delay betweenpresenting a write address to the memory device and the correspondingdata for the write, thus eliminating dead cycles on either the addressbus or the data bus.

FIG. 38 is a schematic diagram of the latch timing circuit forgenerating the major timing signals which control the sense and restoretiming of the bit line sense amplifiers.

FIG. 38A is a waveform diagram which illustrates the waveforms for theinternal nodes of the latch timing sense amplifier shown in FIG. 38.

FIG. 39 is a conceptual diagram of the timing setting control circuitwhich generates a signal for selecting one of several possible latchtiming settings, and which signal may be permanently modified by laserfusing to alter the default timing setting, and may also be temporarilymodified, either before or after laser fusing, by electrical testsignals to alter the timing setting.

FIG. 40 is a timing diagram illustrating the general relationshipbetween major timing signals for an array (read or write) operation forvarious embodiments of a memory array.

FIG. 41 is a block diagram of a portion of a memory bank, illustratingthe row strapping gaps, in which alternating metal1 and metal2 wordlines are each strapped to an associated polysilicon word line segment,which gaps are horizontally aligned with local I/O read amplifier andwrite blocks and located beneath the vertically arranged global inputand output lines generally traversing overhead.

FIG. 42 is a layout diagram of a portion of a memory bank, illustratingthe row strapping gaps depicted in FIG. 41, and which also shows the bitline cross-over structures.

FIG. 43 is a schematic diagram of another embodiment of a column decodearrangement for coupling a selected sense amplifier through a pair oflocal I/O lines to a pair of global output lines when reading, and forcoupling a pair of global input lines through the pair of local I/Olines to the selected sense amplifier when waiting, with the even columnaddresses selecting a sense amplifier below the array block, and the oddcolumn addresses selecting a sense amplifier above the array block, bothof which are coupled to the same set of global input/global outputlines.

In the drawings, depicted elements are not necessarily drawn to scale,and like or similar elements may be designated by the same referencenumeral throughout the several views.

DETAILED DESCRIPTION

Referring now to FIG. 1, a memory circuit 100 is shown whichincorporates various features of the present invention, and whichprovides an exemplary embodiment for describing the features andadvantages of the present invention. A memory array 102 includes, forthis embodiment, a logical size of 4096 rows by 2304 columns, and whichis arranged logically as 4096 rows by 32 columns for each bit within a72-bit double word. Row and column redundancy are also incorporated intoarray 102 which increases the actual number of physical rows andcolumns. A row decoder 104 receives row address information from rowcircuits 106 to decode a selected row (i.e., word line) within array102. Likewise, a column decoder 108 receives column address informationfrom column circuits 110 to decode 72 selected columns (i.e., bit lines)within the array 102. Either 36 or 72 bits are read from (or writteninto) the respective memory cells located at the intersections of theselected word line and the selected bit lines, and conveyed to (or from)36 external terminals 116 through a 72-to-36 bit multiplexer 109(selected by the least significant column address bit) and an I/Ocircuit 114.

A control block 112 receives an external clock signal CLOCK and avariety of other control signals, including a read/write control R/W#,an advance/load control ADV/LOAD #, a chip enable CE, and a clock enableCLKEN. A synchronization circuit 118, such as a phase-locked loop ordelay-locked loop, affords synchronizing internal control signals withthe external clock signal CLOCK. In the preferred embodiment the core ofthe memory device operates at a nominal VDD of 2.5 volts, the I/Osection operates at a nominal VDD of 3.3 volts, and portions of thearray (and related control signals) are driven to aninternally-generated and regulated 4.0 volt level. Various embodimentsof these circuits and their respective features and advantages aredescribed in detail below.

Referring now to FIG. 2, a portion of memory array 102 is illustratedand shows a plurality of sub-arrays or array blocks ARRAY.0, ARRAY.1,ARRAY.2, ARRAY.3, etc. each preferably having, for this embodiment, 256word lines (running horizontally but not shown) and having a pluralityof bit lines configured in a folded-bit-line arrangement. As usedherein, an “array block” is a two-dimensional group of memory cellswhich may contain bit line cross-overs and word line straps (and relatedlayout structures), but no other transistor circuitry. (Depending uponthe capacitance and the resistance of each bit line, then each of theplurality of sub-arrays or array blocks may have other numbers of wordlines, such as 128 word lines, as is discussed in greater detail below.)A plurality of sense amplifiers is located within each of the holesHOLE.0.1, HOLE.1.2, HOLE.2.3 between the array blocks, and within thehole HOLE.0 located above the top array block ARRAY.0. Each senseamplifier is selectable to sense and restore either a bit line pair fromthe array block above the sense amplifier, or a bit line pair from thearray block below the sense amplifier. For example, sense amplifier 122may be configured to sense and restore the bit line pair BL, BLB withinarray block ARRAY.0 if the selected word line is located within arrayblock ARRAY.0, or alternatively may be configured to sense and restorethe bit line pair BL, BLB within array block ARRAY.1 if the selectedword line is located within array block ARRAY.1. Every other bit linepair (e.g., the odd-numbered pairs) within an array block is connectableto a sense amplifier located in the hole above the array block, with theremaining every other bit line pairs (e.g., the even-numbered pairs)within the same array block connectable to a sense amplifier located inthe hole below the array block. Such an alternating sense amplifierarrangement allows the number of sense amplifiers within a hole to beequal to half the number of bit line pairs within each of the arrayblocks. Consequently, the layout of each sense amplifier need only matchthe pitch of two pairs of bit lines rather than one pair.

Sense amplifier 122 includes a first pair of array select transistors126 which, if enabled by a suitable voltage level on an array selectsignal ASU.0 (“array select up, array block 0”), connect the true andcomplement bit lines BL, BLB within array block ARRAY.0 to respectivetrue and complement sense amplifier nodes SA, SAB. The sense amplifier122 further includes a second pair of array select transistors 128which, if enabled by a suitable level on an array select signal ASD.1(“array select down, array block 1”), connect the true and complementbit lines BL, BLB within array block ARRAY.1 to the same respectivesense amplifier nodes SA, SAB within the sense amplifier 122.

Separate array select signals run (through each hole and parallel to theword lines) at the top and at the bottom of each array block. Betweenactive cycles, each array select line is preferably held high (e.g., atthe positive supply voltage VDD) to assist in equilibrating the bitlines. Then, when an active row cycle starts, high-order row addressesare decoded to determine which array block is to be selected. The arrayselect signals at both the top and bottom of the selected array blockremain logically high to provide a path between each bit line pair BL,BLB and corresponding sense amplifier internal nodes SA, SAB. Therespective array select line on the deselected side of each of the tworows of selected sense amplifiers is preferably brought low before theselected word line is driven high, to isolate the respective internalsense amplifier nodes SA, SAB from the capacitance of respective bitlines BL, BLB located within the adjacent but deselected array block.All other array select lines within the memory array 102 preferablyremain high since no other word line is driven active, nor are any othersense amplifiers strobed.

A particular example may provide additional clarity to such a memoryorganization and its operation. If, for example, a selected word linelies within array block ARRAY.1 the array select signals ASD.1 and ASU.1at both the respective top and bottom of array block ARRAY.1 remainhigh, thus coupling each bit line pair within array block ARRAY.1 to acorresponding sense amplifier within either hole HOLE.0.1 or HOLE.1.2.The array select lines ASU.0 and ASD.2 on the deselected side of each ofthe two rows of selected sense amplifiers (i.e., those sense amplifierslocated in holes HOLE.0.1 and HOLE.1.2) are brought low before theselected word line is driven high, to isolate the respective internalsense amplifier nodes SA, SAB from the capacitance of respective bitlines BL, BLB located within deselected array blocks ARRAY.0 andARRAY.2. All other array select lines within the memory array 102 (e.g.,ASD.0, ASU.2, ASD.3, etc.) except these four just described remain highto minimize power dissipation and to provide continuous bit line andsense amplifier equilibration to deselected array blocks (along withother equilibration transistors described below).

Similarly, if a selected word line falls within array block ARRAY.2, thearray select signals ASD.2 and ASU.2 at both the respective top andbottom of array block ARRAY.2 remain high, thus coupling each bit linepair within array block ARRAY.2 to a corresponding sense amplifierwithin either hole HOLE.1.2 or HOLE.2.3. The array select lines ASU.1and ASD.3 on the deselected side of each of the two rows of selectedsense amplifiers are brought low before the selected word line is drivenhigh, to isolate the respective internal sense amplifier nodes SA, SABfrom the capacitance of respective bit lines located within deselectedarray blocks ARRAY.1 and ARRAY.3. All other array select lines withinthe memory array 102 (e.g., ASD.0, ASU.0, ASD.1) except these four justdescribed remain high to minimize power dissipation and to providecontinuous bit line and sense amplifier equilibration to deselectedarray blocks.

Further details of various embodiments of the sense amplifiers and arrayselect signals, including detailed timing and voltage levels, aredescribed in greater detail below.

While the above description related to FIG. 2 illustrates a generalorganization of an exemplary memory array 102 into array blocks and thesharing (i.e., multiplexing) of sense amplifiers between bit line pairsfrom separate array blocks, it provides little indication of how columnsare selected and how each sense amplifier output is steered to I/Ocircuits external to the memory array 102. FIG. 3 is a simplifiedelectrical schematic diagram which illustrates for certain embodimentsthe basic functionality of the memory array 102, and particularlyillustrates the use of a pair of bi-directional local I/O lines LIO,LIOB to couple differential data from the two internal nodes of aselected one of several neighboring bit line sense amplifiers to a localamplifier 174, which then drives a pair of global output lines GOUT,GOUTB. The pair of local I/O lines LIO, LIOB runs horizontally (i.e.,parallel to the word lines) within an array hole and services severalneighboring bit line sense amplifiers located within the same arrayhole, preferably from 4 to 16 sense amplifiers. The global output linesGOUT, GOUTB run vertically (i.e., parallel to the bit lines) and extendthe full length of the memory array. A separate local amplifier (e g.,amplifier 174) is used for each pair of local I/O lines LIO, LIOBlocated within each array hole, and is each selectable (by decodingarray select addresses and other timing information described below) todrive a corresponding pair of complementary global output lines GOUT,GOUTB. Each pair of global output lines GOUT, GOUTB includes a pair ofstatic P-channel load transistors 191, 192 (or other resistive means)and is sensed by a GOUT amplifier 193 (various embodiments of which aredescribed below) to generate an output signal on a global data bus (hereshown implemented as differential global data bus lines GDB, GDBB).

A corresponding pair of vertical global input lines GIN, GINB is used tosteer write data into a selected memory cell. The pair of global inputlines GIN, GINB is momentarily driven by a GIN amplifier 190 with a datasignal received from the complementary global data bus (GDB, GDBB),which are then coupled by a pair of transistors 180, 181 to the pair oflocal I/O lines LIO, LIOB, and which is then coupled to a selected senseamplifier by column select transistors 178, 176. Simplified nomenclatureis used in FIG. 3 for an arbitrary array block, to aid in clarity ofdescription. Subsequent figures introduce circuit schematics which addfull array block decoding and timing necessary to support a memory array102 having a large number of array blocks, as is described more fullybelow.

Having briefly set forth thus far the basic I/O structure of the memoryarray 102, a more detailed description follows below in the context of abasic read cycle, followed by a description of a basic write cycle. Across-coupled CMOS sense amplifier 142 is multiplexed, as describedabove, to sense either a bit line pair BLU, BLBU (i.e., “bit line up”and “bit line bar up”) within the array block located above the senseamplifier 142, or to sense a bit line pair BLD, BLBD (i.e., “bit linedown” and “bit line bar down”) within the array block located below thesense amplifier 142. To select the upper bit line pair BLU, BLBU, anupper array select signal ASU is left high while a lower array selectsignal ASD is brought low, as described above in reference to FIG. 2.Array select transistors 160, 162 remain on and couple the upper bitline pair BLU, BLBU to respective sense amplifier nodes SA, SAB whilearray select transistors 164, 166 are turned off to isolate the senseamplifier nodes SA, SAB from the lower bit line pair BLD, BLBD.Conversely, to select the lower bit line pair BLD, BLBD, the lower arrayselect signal ASD is left high while the upper array select signal ASUis brought low to isolate the unselected bit line pair within the upperarray block from the sense amplifier internal nodes SA, SAB. Preferablythe array select signals which remain logically high (to gate theselected bit lines to the respective sense amplifier) are actually“boosted” or driven to a voltage above the VDD level to provide for alower impedance path, particularly during restoration of the high levelbit line voltage, as is discussed in greater detail below. If boostedarray select levels are employed, then the array select transistors(e.g., array select transistors 160, 162) may be preferably fabricatedusing a high voltage transistor structure if one is available in thesemiconductor process being used. However, as is described in greaterdetail below, the array select signals may be boosted to a VPP voltagewhich, for reliability reasons, is a regulated, fixed “safe” voltagelevel referenced to VSS, but which is typically well above VDD fortypical operating conditions. If two different transistor structures(e.g., normal and high voltage) are not available, all the transistors,including the memory cell access transistors, are designed to workreliably at the boosted voltage.

The general column organization of this embodiment of the memory array102 may be illustrated by considering a memory cycle in which theselected word line falls within, for example, the array block locatedbelow the sense amplifier 142 in FIG. 3. A word line WL is shown which,when driven active, couples a memory cell 150 to the complement bit lineBLBD. The memory cell 150 includes NMOS access transistor 146 whichcouples one terminal of a memory cell capacitor 148 to the bit lineBLBD, while the other terminal of memory cell capacitor 148 is connectedto memory cell plate 152. As is common practice with all DRAMs, half ofthe memory cells associated with a given bit line pair connect to thetrue bit line, and the remaining half connect to the complement bitline. A portion of the row decoder for the word line WL includes rowdriver 144, which is powered from a VPP level which is higher than thenormal VDD level used throughout the majority of the circuit. This VPPvoltage is preferably internally generated and regulated (using abandgap reference voltage) to a substantially constant value of 4.0volts relative to VSS (for a memory device designed to work using anominal VDD level of 2.5 volts) independent of semiconductor processcorner, temperature, and the particular VDD level. NMOS accesstransistor 146 is preferably fabricated using a high voltage transistorstructure, if available, as is indicated by the adjoining notation “*”in FIG. 3. If no special high voltage transistor structure is available,all transistors are designed (based upon reliability considerations) totolerate a VPP voltage of, for example, 3.6 to 4.0 volts, even thoughthe circuit is intended to operate at a VDD of nominally only 2.5 volts.

After a previous active cycle (and consequently while all the row linesare held inactive low) the complementary bit lines BLD, BLBD areequilibrated to each other by equilibration transistor 156 which isgated by a self-timed pulsed equilibrate signal BLEQD and whichtransistor is located at the “near end” of the bit line pair (next tothe array select transistors 164, 166). Because the sense amplifier 142restores one bit line to the low supply voltage level (i.e., VSS orground) and the other bit line to a high voltage level, the bit linessubsequently establish, after sharing their charge, a bit lineequilibration voltage that is approximately one-half of the high write(i.e., restore) level. The sense amplifier nodes SA, SAB areequilibrated to each other by transistor 182 which is gated by a senseamplifier equilibrate signal SAEQ_LEVEL which remains high until thenext active cycle begins in which sense amplifier 142 is selected. Thesense amplifier nodes SA,SAB are further equilibrated to a common (i.e.,shared) bit line equilibrate node VBLEQ by transistors 154, 158 whichare gated by a self-timed pulsed sense amplifier equilibrate signalSAEQ_PULSE, to establish the bit line equilibration voltage of thecollective high capacitance of all the bit line pairs on the relativelylow capacitance of the common bit line equilibrate node VBLEQ. TheSAEQ_PULSE signal is self-timed and remains high for the same durationas the other pulsed equilibrate signals, which is for a sufficient timeto adequately equilibrate the sense amplifier nodes SA, SAB, and toestablish the bit line equilibration voltage faithfully onto the commonbit line equilibrate node VBLEQ, but the SAEQ_PULSE signal is thenautomatically brought low without waiting for a new cycle to begin. (Theequilibration transistor 182 may alternatively be implemented as twoseparate, parallel transistors: one gated by the sense amplifierequilibrate signal SAEQ_LEVEL, and the other gated by the senseamplifier equilibrate signal SAEQ_PULSE.) The lower bit line pair BLD,BLBD is also equilibrated at the “far end” of the bit lines (i.e., theend most distant from its sense amplifier) by equilibrate transistor168, also gated by a decoded equilibrate signal having the same timingas pulsed equilibrate signal BLEQD (represented by a dashed line in FIG.3). By equilibrating the bit lines from both ends, the equilibrationtime is reduced by almost a factor of four as compared to equilibratingfrom only one end of the bit lines (assuming the “on” resistance of eachequilibrate transistor is small compared to the bit line resistance,represented schematically as parasitic bit line resistors 197).

The upper bit line pair BLU, BLBU is similarly equilibrated byequilibration transistor 170 which is gated by a self-timed pulsedequilibrate signal BLEQU and which transistor is preferably located atthe near end of the upper bit line pair (next to the array selecttransistors 160, 162). The upper bit line pair is also equilibrated atits far end by an equilibrate transistor (not shown) analogous totransistor 18, also gated by a decoded equilibrate signal having thesame timing as pulsed equilibrate signal BLEQU.

After the previous active cycle, both array select signals ASU, ASD aredriven to VDD (if not already at such a voltage). The additional senseamplifier equilibration transistor 182, which is gated by equilibratesignal SAEQ_LEVEL, directly equilibrates the internal sense amplifiernodes SA, SAB together. Consequently, both the true and complement upperbit lines BLU, BLBU, the true and complement sense amplifier nodes SA,SAB, and the true and complement lower bit lines BLD, BLBD are allequilibrated together by transistor 182 until the next cycle using senseamplifier 142. These lines, when initially equilibrated, jointsestablish the bit line equilibration voltage onto node VBLEQ and whichvoltage, for this embodiment, is equal to approximately 1.0 volt, as isfurther described below.

The local I/O lines LIO, LIOB are also equilibrated to the bit lineequilibration voltage by transistors 185, 186. A local I/O equilibrationsignal LIOEQ provided to the gate terminals of both transistors 185, 186is driven high between active cycles and remains high even during activecycles whenever the particular local I/O line pair is deselected (i.e.,its hole is deselected). Between active cycles, strobed amplifier 174 isinactive, allowing the pair of global output lines GOUT, GOUTB to beboth driven to VDD by respective static load transistors 191, 192.

When an active cycle starts in the lower array, the equilibrate signalSAEQ_LEVEL is brought low to de-couple the two internal sense amplifiernodes SA, SAB from each other. The other pulsed equilibrate signals(e.g., the SAEQ_PULSE and BLEQD signals in all the array blocks) arealready low (for long cycle times) or are concurrently being brought low(for minimal cycle times) in order to reduce the capacitance of various“equilibrate” signals which must be brought low at the start of anactive cycle before the selected word line may be driven high. Moreover,when an active cycle starts in the lower array, the ASU signal is alsobrought low to de-couple the internal sense amplifier nodes SA, SAB fromthe bit line pair BLU, BLBU in the de-selected array block located abovethe sense amplifier 142. Upper-order row addresses (i.e., array blockselect addresses) are decoded by the respective circuits which generatethe various sense amplifier equilibrate signals and array select signalsto determine, based upon which array block is selected, which of thesignals to bring low. The LIOEQ signal is also brought low to de-couplethe local I/O lines LIO, LIOB from each other, again based uponupper-order row address information (as well as internally-generatedtiming information).

As soon as the appropriate equilibrate and array select signals aresufficiently low, the selected word line WL is driven high. Thesewaveforms may actually overlap slightly, as the voltage of theequilibrate and array select signals must preferably be belowapproximately one N-channel threshold voltage above the bit lineequilibrate level before the voltage of the selected word line WLreaches the threshold voltage of the N-channel access transistor 146.Because the word line driver 144 is implemented using a very largeoutput pull-up transistor, and is powered by an internally generated andregulated 4.0 volt supply voltage (VPP) rather than by VDD, the selectedword line WL rises smoothly and quickly to a 4.0 volt high level tofacilitate restoration of a high level into a memory cell. As theselected word line WL rises, of course, a differential signal isgenerated between the differentials bit lines BLD, BLBD, depending onwhether the stored data within the selected memory cell 150 was earlierwritten to either a high voltage (e.g., about 2.0 volts) or a lowvoltage (about 0 volts). Because the array select transistors 164, 166remain conductive (because array select signal ASD remains high) thedifferential signal on the bit lines BLD, BLBD is coupled onto theinternal sense amplifier nodes SA, SAB.

At substantially the same time as the selected word line WL is drivenhigh, and particularly of note before the sense amplifier 142 isstrobed, the selected column decode signal COLSEL is also driven high,in this case to a VDD level. Consequently, the developing signal on thesense amplifier nodes SA, SAB is also coupled to the very short, verylow capacitance local I/O lines LIO, LIOB before sensing of the senseamplifier 142. Moreover, at substantially the same time and still beforesensing of the sense amplifier 142, the local I/O amplifier 174 isenabled (as determined by decoded address information combined withinternal timing signals, not yet shown) and the developing signal on thelocal I/O lines LIO, LIOB is buffered onto the global output lines GOUT,GOUTB. As soon as a sufficient signal is developed on the internal senseamplifier nodes SA, SAB, the bit line sense amplifier 142 is strobedsimultaneously by strobe enable signal SE and strobe enable bar signalSEB. When strobed, the sense amplifier 142 restores a low level on thebit line having a lower sense voltage and restores a high level on thebit line having a higher sense voltage, as is well known in the art. Therelative timing of these strobe enable signals will be described ingreater below, but for ease of description the strobing of the senseamplifier 142 may at this point be assumed to restore adequate high andlow levels onto the respective bit lines connected thereto. With theselected word line WL still high, the selected memory cell voltage isrestored as well.

In a traditional DRAM, the signal from the selected memory cell isdeveloped in and strobed by the sense amplifier before any column decodesignal is driven which couples either the selected bit lines and/orsense amplifiers to any type of I/O lines. One reason this is done is toensure maximum signal in the sense amplifiers before sensing and toreduce any interference that the column select signals might impart uponproper sensing of the data from the selected memory cell. Connection ofa very large capacitance global I/O line to a low capacitance bit linebefore sensing would severely attenuate the signal to be sensed. Anotherreason this is done include I/O lines which are advantageouslyequilibrated to a much different voltage than are bit lines.

In the embodiment shown in FIG. 3, the selected column decode signal isdriven before sensing. This counter-intuitive technique slightlyattenuates the signal available to the sense amplifiers due to the smallincreased capacitive loading of a very short, low capacitance local I/Oline (e.g., LIOB) onto the combined capacitance of a bit line senseamplifier node (e.g., SAB) and a bit line (e.g., BLBD), but itadvantageously affords an opportunity to start developing signal ontothe global output lines GOUT, GOUTB even before sensing has occurred.This provides a tremendous read access time advantage compared towaiting for sensing before driving selected column decode signals. Itshould be appreciated that the sense amplifier nodes SA, SAB must not bedisturbed or mis-equilibrated when the selected column decode signalCOLSEL is driven high. Because the local I/O lines are precharged to thesame voltage as are the bit lines and sense amplifier nodes (e.g., thebit line equilibrate voltage established on node VBLEQ), the common modevoltage of the bit line sense amplifier nodes nominally is unchangedwhen the selected column decode signal COLSEL is driven high because theequilibrate voltage is the same. Careful attention to balanced layoutresults in reducing any differential capacitive coupling through columnselect transistors 176, 178, which preserves the equilibration of thesense amplifier nodes SA, SAB.

The global output lines GOUT, GOUTB run the entire vertical length ofthe memory array 102, as described above. At substantially the same timeas the selected word line WL and the selected column decode signalCOLSEL are driven high, in a read cycle the decoded read signalHOLE_READ is driven high (as determined by decoded array select addressinformation combined with internal timing signals, not yet shown), alsobefore sensing. Thus, as the low voltage differential signal develops onthe bit lines BL, BLB, the sense amplifier nodes SA, SAB, and the localI/O lines LIO, LIOB, it is buffered by the selected local I/O senseamplifier 174 and driven onto the differential global output lines GOUT,GOUTB. Load transistors 191, 192 need not be decoded, as the voltage ofthe both global output lines GOUT, GOUTB remain at VDD between activecycles, and during an active read cycle only one of the global outputlines (e.g., GOUT or GOUTB) is briefly driven downward by the selectedlocal I/O sense amplifier 174 to develop a relatively small differentialsignal level between the global output lines GOUT, GOUTB. Additionaldetails of the local I/O sense amplifier 174 and the GOUT amplifier 193,as well as complete data path details from the memory array to theexternal connections of the exemplary embodiment, are described furtherbelow. An alternate embodiment may, include several decoded GOUT loadresistors along the length of a given global output line, which aredecoded such that only the load resistor located adjacent to or at leastphysically nearest the selected local I/O sense amplifier is enabledduring a read cycle, to minimize the voltage drop on the global outputlines between the local I/O sense amplifier and the load resistor.However, this also adds significant complexity in control circuits todecode the load resistors appropriately. Other plausible embodimentsinclude a decoded load resistor at both the ¼ and ¾ points along theglobal output lines, a non-decoded load resistor at both the top andbottom of the memory bank, or a non-decoded load resistor at just theend of the memory bank having the GOUT sense amplifier 193 (as shown inFIG. 3).

In a write operation, both the data to be written and the write addressare known even before the start of the write operation. In the exemplaryembodiment, this occurs because a write queue “buffers up” several writecycles before any data is actually written into the array. Additionaldetails of the write queue are described in greater detail below.Nonetheless, to understand the basic write operation of the simplifiedcircuitry as shown in FIG. 3, one may assume the data is known to thememory before the start of the write cycle which actually “writes” thedata into the selected memory cells. More specifically, the data whichis to be next written is already placed upon the global input lines GIN,GINB before an active cycle starts. Even more specifically, adifferential voltage is driven onto the global input lines GIN, GINBduring the latter portion of bit line sensing and the early portion ofbit line equilibration so that the global input lines are not movingduring the latter portion of bit line equilibration and through theearly portion of bit line sensing when capacitive coupling to anunderlying bit line or sense amplifier node could disturb bit linesensing.

At substantially the same time as the selected word line WL and theselected column decode signal COLSEL are driven high, in a write cyclethe decoded write signal HOLE_WRITE is driven high (as determined bydecoded array select address information combined with internal timingsignals, not yet shown), also before sensing. This couples the lowvoltage differential write data signal already present on the globalinput lines GIN, GINB onto the local I/O lines LIO, LIOB by way oftransistors 180, 181. Since the selected column decode signal COLSEL hasalso been driven (or is being driven) high, the write data signal isalso coupled onto the internal sense amplifier nodes SA, SAB, where itis combined with the signal otherwise developed by the selected memorycell. The magnitude of the differential write data signal on the globalinput lines GIN, GINB is preferably chosen to nominally impart a voltageon each bit line which is no larger than might otherwise occur during aread cycle. Due to the much larger capacitance of the global input linesGIN, GINB compared to the sum of the respective capacitance of the bitlines BLD, BLBD, the bit line sense amplifier nodes SA, SAB, and thelocal I/O lines LIO, LIOB, the resulting voltage developed in the bitline sense amplifier 142 is almost as large as the differential writedata signal itself. At the same time, the signal developed from theselected memory cell is negligible, as most of this charge ends up onthe very large capacitance of the global input line. That is, the globalinput line substantially “swallows” the signal otherwise developed fromreading the selected memory cell.

On the other hand, the magnitude of such a write data signal should belimited in order to reduce the magnitude of unwanted coupling ontoneighboring sense amplifier nodes SA, SAB that are not being written(and whose restore operation must not be disturbed) when the senseamplifier nodes and bit lines to be written are driven with the writedata signal. As a point of reference, during a read cycle one bit line(of a selected bit line pair) is either brought up by about 100 mV whenreading a “high” or brought down by about 100 mV when reading a “low”,while the other bit line of the selected bit line pair remainssubstantially at the bit line equilibrate voltage. In other words, themaximum voltage perturbation of a given bit line which might disturb aneighbor is either +100 mV or −100 mV. The magnitude of the write datasignal on each global input line GIN, GINB is preferably chosen tonominally be equal to the expected read signal from the memory cell,that is, 100 mV above or below the bit line equilibration voltage. Thusthe write data signal coupled to the selected bit line sense amplifierand bit line pair, will not disturb a neighboring bit line senseamplifier more than a read cycle would disturb that neighbor.

One global input line is driven to 100 mV above the bit line equilibratevoltage, while the other global input line is driven to 100 mV below thebit line equilibrate voltage. As stated above, due to the much largercapacitance of the global input lines, the resulting voltage on theselected bit line sense amplifier nodes and likewise on the selected bitline pair is only slightly attenuated from these voltage levels.Consequently, although a differential signal of 200 mV is developed inthe bit line sense amplifier (twice that developed in a read cycle),each sense amplifier node and each bit line is driven by only 100 mV,which is no worse than the perturbation to a neighbor during a readcycle.

Because of this timing, a small differential voltage representing thewrite data is actually driven into the sense amplifier before sensing.As stated above, the magnitude of this small differential voltage may bepreferably chosen to be comparable to, but somewhat larger than, theexpected signal otherwise developed during reading. The sense amplifier142 then latches the write data when strobed, rather than latching theread data from the memory cell. This technique has several significantadvantages. First, as mentioned above, it does not disturb theneighboring bit line sense amplifiers. Second, it is fast! The bit linesense amplifiers do not first sense “old” data and then overwrite itwith new data, which would require significant additional time. Rather,they initially sense the data to be written. Third, the global inputlines only move +/−100 mV during each write cycle. Since there is a36-bit wide data path and each global input line has over 1 pF ofcapacitance, considerable power is saved by only moving these wires by100 mV. Fourth, since thle bit line sense amplifier itself restores thelevels onto the bit lines, the write levels into a memory cell aresubstantially identical to the restored level achieved after a read orrefresh cycle. This is particularly advantageous in ensuring the highlevel during write matches the high level during read (which isextremely beneficial for reducing functional test times). Moreover, auniform written high level is achieved, particularly when not writingall the way to VDD, irrespective of how many bits are written in a givencycle (because of the byte write capability). A fifth advantage is foundin the lack of any requirement that the write circuits must over-power alatched sense amplifier. If required, such circuits would be physicallylarger and consume even more power, and would require a write path otherthan the simple NMOS-only column select transistors, which alone couldnot drive the sense amplifier nodes all the way up the write restorelevel.

The sense amplifier latches either the data read from a selected memorycell, or data to be written to the selected memory cell. The senseamplifier is then responsible for restoring the proper high and lowlevels onto the complementary bit lines, and consequently into theselected memory cell. The selected word line WL is driven in an activecycle to an internally regulated VPP voltage which, for nominal powersupply voltage, is above the VDD level. In other words, for nominalpower supply values, the selected word line is boosted above VDD inorder to reduce the effective resistance of the access transistor 146,thereby restoring a high level into the selected memory cell 150 in lesstime. For most process corners (especially aggravated by high threshold,low VDD, and cold temperature), the boosted word line level alsoovercomes the potential loss of the access transistor threshold voltagewhich would otherwise limit the high level which may be written intomemory cell 150.

For much the same reasons as for the word line, the array selecttransistors 164, 166 could also limit either the ultimate voltage levelor at least impact the delay of restoring a high level onto theappropriate true or complement bit line. Consequently, the array selectsignal ASD is also boosted to the VPP level during the restore portionof the active cycle. The array select signal ASD is biased at VDDbetween active cycles, as are all the other array select signals. Then,at the start of an active cycle, the array select signals on thedeselected side of the selected sense amplifiers (e.g., array selectsignal ASU) are quickly brought to ground, and the array select signalson the selected side of the two rows of selected sense amplifiers (e.g.,array select signal ASD) are boosted from VDD to VPP to more quicklyallow passage of the read signal from the selected memory celldeveloping on the various bit lines onto the respective internal senseamplifier nodes SA, SAB, and then to much more quickly allow passage ofthe “latched signal” developing on the sense amplifier nodes duringsensing (particularly a high-going restore level) back onto therespective bit lines.

In the exemplary embodiment shown, the bit line equilibrate voltage isapproximately 1.0 volt, and the written/restored high level isapproximately 2.0 volts (which is described in additional detail below).Even at a low VDD of 2.1 volts, a VDD level on the “selected” arrayselect signal is most likely sufficient to couple the true andcomplement bit line voltages, as the approximately 100 mV differentialsignal develops from reading a selected memory cell, onto the true andcomplement sense amplifier nodes SA, SAB. But such a 2.1 volt level onthe gates of array select transistors 164, 166 would, for this example,limit the level developed of the high-going bit line when the senseamplifier drives the high-going sense amplifier node to 2.0 volts.Consequently, during the time signal is developing on the bit lines andinto the sense amplifier, the “selected” array select signal is boostedto the VPiP level (nominally 4.0 volts) by circuitry which need not beparticularly fast, as long as the voltage on the array select signal isdriven to VPP fairly early in the sensing process.

The memory cell plate voltage PLATE (node 152) is generated by asample-and-hold amplifier 196 to match the bit line equilibrationvoltage, which is sampled during the precharge pulse (of signalSAEQ_PULSE) by transistor 194 onto capacitor 195.

Referring now to FIG. 4, an arrangement of local I/O lines is describedwhich shows a pair of local I/O lines which are “broken” in the middle(to reduce capacitance), thus forming a left complementary local I/Opair and a right complementary local I/O pair. The left and right localI/O line pairs are then connected to share a single local I/O readamplifier/write block 202. Thirty-two physical bit line pairs from aportion of, for example, array block ARRAY.2 are shown. As waspreviously described in relation to FIG. 2, half of the bit line pairsare connected to bit line sense amplifiers (indicated as BLSA) locatedin the hole HOLE.1.2 above the array block ARRAY.2, while the other halfof the bit line pairs are connected to bit line sense amplifiers locatedin the hole HOLE.2.3 below the array block ARRAY.2. This circuitry maybe horizontally repeated 36 times across each memory bank.

Of the sixteen physical bit line pairs on the left side of FIG. 4(indicated as BLP.0, BLP.1, BLP.2, . . . BLP.15), the eighteven-numbered bit line pairs BLP.0, BLP.2, . . . BLP.14 are connected torespective bit line sense amplifiers SA.0, SA.2, . . . SA.14 at the topof the array block ARRAY.2 (i.e., within the hole HOLE.1.2). Each ofthese eight even-numbered bit line sense amplifiers is then connected toa corresponding left-hand pair of local I/O lines LIO_L, LIOB_L. Of thesixteen physical bit line pairs on the right side of FIG. 4 (indicatedas BLP.16, BLP.17, BLP.18 . . . BLP.31), the eight even-numbered bitline pairs BLP.16, BLP.18, . . . BLP.30 are connected to respective bitline sense amplifiers SA.16, SA.18, . . . SA.30 at the top of the arrayblock ARRAY.2. Each of these eight even-numbered bit line senseamplifiers is then connected to a corresponding right-hand pair of localI/O lines LIO_R, LIOB_R.

Both the left local I/O pair LIO_L, LIOB_L and the right local I/O pairLIO_R, LIOB_R are brought to a local I/O read amplifier and write block202, which is selectable during an active cycle to couple either theleft or the right local I/O pair to an associated set of global I/Olines (e.g., “global I/O lines GI/O.2,” which includes a differentialglobal input pair GIN.2, GINB.2, and a differential global outut pairGOUT.2, GOUTB.2). Note that the complementary global output lines GOUT.2and GOUTB.2 are not routed adjacent to each other, but are insteadseparated by one of the global input lines (e.g., GINB.2). Thissignificantly reduces the coupling capacitance between the true andcomplement global output lines GOUT.2 and GOUTB.2, and since the globalinput lines GIN.2, GINB.2 are not moving (i.e., they are “quiet”) whenthe read signal is developing on the global output lines, the delay inthe read path is reduced.

A group of eight pre-decoded column select lines COLSEL.X runs througheach hole (e.g., hole HOLE.1.2, HOLE.2.3, etc.). For ease ofintroduction, assume a group of two pre-decoded left/right select linesLEFT, RIGHT also runs through each hole as shown in FIG. 4 for holeHOLE.1.2 and hole HOLE.2.3. During an active cycle, one of the eight bitline sense amplifiers connected to the left local I/O pair LIO_L, LIOB_Lis selected by the active one of the eight pre-decoded column selectlines COLSEL.X in accordance with a portion of the column address of thegiven active cycle. Likewise, one of the eight bit line sense amplifiersconnected to the right local I/O pair LIO_R, LIOB_R is also selected bythe same active one of the eight pre-decoded column select linesCOLSEL.X. The active LEFT or RIGHT select line (which is also derivedfrom a portion of the column address of the given active cycle) thensteers the local I/O read amplifier and write block 202 appropriately toeither the left local I/O pair LIO_L, LIOB_L or the right local I/O pairLIO_R, LIOB_R. The local I/O read amplifier and write block 202 buffersthe signal from the selected local I/O pair onto the associated globaloutput pair GOUT.2, GOUTB.2 (when enabled during a read cycle by a READsignal routed through the hole HOLE.1.2), or drives the data signal fromthe associated global input pair GIN.2, GINB.2 onto the selected localI/O pair (when enabled during a write cycle by a WRITE signal alsorouted through the hole HOLE.1.2).

In a similar fashion, the eight odd-numbered bit line pairs BLP.1,BLP.3, . . . BLP.15 on the left side of FIG. 4 are connected torespective bit line sense amplifiers SA.1, SA.3, . . . SA.15 in holeHOLE.2.3 below the array block ARRAY.2. Each of these eight odd-numberedbit line sense amplifiers is then connected to a corresponding left-handpair of local I/O lines LIO_L, LIOB_L. Of the sixteen physical bit linepairs on the right side of FIG. 4 (indicated as BLP.16, BLP.17, BLP.18 .. . BLP.31), the eight odd-numbered bit line pairs BLP.17, BLP.19, . . .BLP.31 are connected to respective bit line sense amplifiers SA.17,SA.19, . . . SA.31 at the bottom of the array block ARRAY.2. Each ofthese eight odd-numbered bit line sense amplifiers is then connected toa corresponding right-hand pair of local I/O lines LIO_R, LIOB_R.

Both the left local I/O pair LIO_L, LIOB_L and the right local I/O pairLIO_R, LIOB_R are brought to a local I/O read amplifier and write block204, which is selectable during an active cycle to couple either theleft or the right local I/O pair to an associated odd-numbered globalI/O line, in this case, global I/O line GI/O.3 (which includes adifferential global input pair GIN.3, GINB.3, interspersed with adifferential global output pair GOUT.3, GOUTB.3).

As stated above, a group of eight pre-decoded column select linesCOLSEL.X (individually identified as COLSEL.0, COLSEL.1, . . . COLSEL.7)also runs through hole HOLE.2.3. For ease of description, again assume agroup of two pre-decoded left/right select lines LEFT, RIGHT also runsthrough hole HOLE.2.3, as shown in FIG. 4. During an active cycle, oneof the eight odd-numbered bit line sense amplifiers connected to theleft local I/O pair LIO_L, LIOB_L is selected by the active one of theeight pre-decoded column select lines COLSEL.X in accordance with aportion of the column address of the given active cycle. Likewise, oneof the eight odd-numbered bit line sense amplifiers connected to theright local I/O pair LIO_R, LIOB_R is also selected by the same activeone of the eight pre-decoded column select lines COLSEL.X. The activeLEFT or RIGHT select line (which is also derived from a portion of thecolumn address of the given active cycle) then steers the local I/O readamplifier and write block 204 appropriately to either the left local I/Opair LIO_L, LIOB_L or the right local I/O pair LIO_R, LIOB_R. The localI/O read amplifier and write block 204 buffers the signal from theselected local I/O pair onto the associated global output pair GOUT.3,GOUTB.3 (when enabled during a read cycle by a READ signal routedthrough the hole HOLE.2.3), or drives the data signal from theassociated global input pair GIN.3, GINB.3 onto the selected local I/Opair (when enabled during a write cycle by a WRITE signal also routedthrough the hole HOLE.2.3). Note that in a single-word (36-bit) read orwrite cycle, the column select and read or write signals in either holeHOLE.1.2 or those in HOLE.2.3 are enabled, based upon the LSB of theentire address field. For a 72-bit double-word read or write cycle,these signals are enabled in both the holes HOLE.1.2 and HOLE.2.3.

A specific example of an active read cycle may be useful to reinforcethe details of this organization. Assume the selected word line (notshown in FIG. 4) falls within array block ARRAY.2 (meaning that arraydecode addresses are decoded to select the array block ARRAY.2, and thatrow decode addresses are decoded to select a word line within arrayblock ARRAY.2 as the active word line). Further assume that columndecode addresses are decoded such that column select line COLSEL.1 andthe left/right select line LEFT in hole HOLE.1.2 are both active. As aresult, bit line sense amplifier SA.2 is selected by column select lineCOLSEL.1 and the signal developed by the selected memory cell on the bitline pair BLP.2 (and subsequently amplified by bit line sense amplifierSA.2) is coupled onto the left local I/O lines LIO_L, LIOB_L within holeHOLE.1.2. At the same time, bit line sense amplifier SA.18 is alsoselected by column select line COLSEL.1 and the signal developed by theselected memory cell on the bit line pair BLP.18 (and also subsequentlyamplified by bit line sense amplifier SA.18) is coupled onto the rightlocal I/O lines LIO_R, LIOB_R within hole HOLE.1.2. Within the local I/Oread amplifier and write block 202, the left local I/O lines LIO_L,LIOB_L are selected (by the active select line LEFT) and coupled to theinput of a read amplifier, and the corresponding read signal developedon the selected local I/O lines is buffered and driven onto theassociated even-numbered global output lines GOUT.2, GOUTB.2. Thenon-selected right local I/O lines LIO_R, LIOB_R are de-coupled from theread amplifier (and thus de-coupled from the global output lines) andare also de-coupled from the global input lines when the left local I/Olines are selected.

At the same time during this exemplary (72-bit double-word) read cycle,odd-numbered bit line sense amplifier SA.3 is also selected by columnselect line COLSEL.1 and the signal developed by the selected memorycell on the bit line pair BLP.3 (and subsequently amplified by bit linesense amplifier SA.3) is coupled onto the left local I/O lines LIO_L,LIOB_L within hole HOLE.2.3. Bit line sense amplifier SA.19 is alsoselected by column select line COLSEL.1 and the signal developed by theselected memory cell on the bit line pair BLP.19 (and subsequentlyamplified by bit line sense amplifier SA.19) is coupled onto the rightlocal I/O lines LIO_R, LIOB_R within hole HOLE.2.3. Within the local I/Oread amplifier and write block 204, the left local I/O lines LIO_L,LIOB_L are selected by the active select line LEFT, and thecorresponding read signal developed on the selected local I/O lines isbuffered and driven onto the associated odd-numbered global output linesGOUT.3, GOUTB.3. As before, the non-selected right local I/O linesLIO_R, LIOB_R are de-coupled from the read amplifier (and thusde-coupled from the global output lines) and are also de-coupled fromthe global input lines when the left local I/O lines are selected.

As described above, during an active cycle when the selected row fallswithin array block ARRAY.2, bit line sense amplifiers are active in theholes immediately above and below the selected array block ARRAY.2,namely, within holes HOLE. 1.2 and HOLE.2.3. For a single-word read orwrite operation, the local I/O lines, column select lines, etc. areactive in either the hole HOLE.1.2 immediately above, or in holeHOLE.2.3 immediately below, the selected array block ARRAY.2.Alternatively, for a double-word read or write operation, the local I/Olines, column select lines, etc. are active in both these holes HOLE.1.2and HOLE.2.3.

For ease of description, during such an active cycle both holes HOLE.1.2and HOLE.2.3 may be considered to be “selected” and all other holes(e.g., holes HOLE.0, HOLE.0.1, HOLE.3.4, HOLE.4.5, . . . HOLE.15) areconsidered “deselected” or “non-selected.” Consequently, the pre-decodedcolumn select lines COLSEL.X and the pre-decoded left/right select linesLEFT and RIGHT in the two selected holes above and/or below the selectedarray block are active. All other column select lines and left/rightselect lines within non-selected holes of the array remain inactive toensure only one local I/O sense amplifier block per global I/O isactive, and to conserve power.

As can be appreciated from FIG. 4, a selected one of sixteeneven-numbered bit line pairs is coupled to an even-numbered global I/Oline, while a selected one of sixteen odd-numbered bit line pairs iscoupled to an odd-numbered global I/O line. To illustrate the lateralrepetition of this array block layout, an adjacent physical bit linepair BLP.32 is shown on the right side of FIG. 4 which is coupled to aneighboring left local I/O pair, and which ultimately is coupled to theadjacent global I/O line (e.g., GIN.4, GINB.4, GOUT.4, GOUTB.4, notshown).

The general structure shown in FIG. 4 is repeated both horizontally andvertically to form the array 102 first introduced in FIG. 1 anddescribed in additional detail in FIG. 2. Referring now to FIG. 5, anorganization is shown which includes two array banks 251, 253, eachrespectively served by a corresponding independent row decoder 252, 254.Within the left bank 251, the structure shown in FIG. 4 is repeatedhorizontally 36 times (not including redundant elements), for a total of72 global I/O lines within the left bank 251. The 36 even-numberedglobal I/O lines (e.g., GI/O.0, GI/O.2 . . . GI/O.70) are brought out atthe bottom of the array bank 251, while the 36 odd-numbered global I/Olines (e.g., GI/O.1, GI/O.3 . . . GI/O.71) essentially “stop” at thebottom of the array bank 251 (other than a “far end” equilibrationcircuit described in detail below). Conversely, the 36 odd-numberedglobal I/O lines GI/O.1, GI/O.3, . . . GI/O.71 are brought out at thetop of the array bank 251, while the 36 even-numbered global I/O linesGI/O.0, GI/O.2, . . . GI/O.70 essentially “stop” at the top of the arraybank 251 (again, other than a “far end” equilibration circuit describedin detail below). As previously stated above, a given “global I/O lineGI/O.X” is a short-hand notation for a group of four physical wireswhich include a differential global input pair GIN.X, GINB.X and adifferential global output pair GOUT.X, GOUTB.X). Within the left bank251, there are 16 different array blocks ARRAY.0, ARRAY.1, . . .ARRAY.15, each array block having 256 word lines (for this exemplaryembodiment). The alternating nature of the global I/O connections withineach hole is indicated by a connecting “dot” in FIG. 5. In particular,one instance of the structure shown in FIG. 4 is indicated in FIG. 5 bythe dashed outline of block 255. Of the two global I/O lines passingthrough block 255, the left-hand global I/O line (e.g., GI/O.2) isconnected to left and right pairs of local I/O lines located above thearray block ARRAY.2 within hole HOLE.1.2, while the right-hand globalI/O line (e.g., GI/O.3) is connected to left and right pairs of localI/O lines located below the array block ARRAY.2 within hole HOLE.2.3.

Within the right bank 253, the structure shown in FIG. 4 is alsorepeated horizontally 36 times, for a total of 72 global I/O lineswithin the right bank 253. The 36 even-numbered global I/O lines (e.g.,GI/O.72, GI/O.74, . . . GI/O.142) are brought out at the bottom of thearray bank 253, while the 36 odd-numbered global I/O lines (e.g.,GI/O.73, GI/O1.75, . . . GI/O.143) essentially “stop” at the bottom ofthe array bank 253 (other than a “far end” equilibration circuitdescribed in detail below). Conversely, the 36 odd-numbered global I/Olines GI/O.73, GI/O.75, . . . GI/O.143 are brought out at the top of thearray bank 253, while the 36 even-numbered global I/O lines GI/O.72,GI/O.74, . . . GI/O.142 essentially “stop” at the top of the array bank253 (again, other than a “far end” equilibration circuit described indetail below). Within the right bank 253, there are 16 different arrayblocks ARRAY.16, ARRAY.17, . . . ARRAY.31, each having 256 word lines(for this exemplary embodiment).

At the bottom of both array banks 251, 253, a multiplexer 256 provides a72-to-18 data selection path for selecting 18 bits of data from the 72even-numbered global I/O lines at the bottom of the array 102, which arethen connected to appropriate data I/O circuitry (not shown in FIG. 4)which is described in detail below. Similarly, at the top of both arraybanks 251, 253, a second multiplexer 257 provides a 72-to-18 dataselection path for selecting 18 bits of data from the 72 odd-numberedglobal I/O lines at the top of the array 102, which are also thenconnected, to similar data I/O circuitry (not shown in FIG. 4).Consequently, the memory array 102 provides a 36-bit data path. Half ofthe data bits (e.g., data bits DATA.0, DATA.1, . . . DATA.17) areavailable at the bottom of the array 102, while the other half of thedata bits (e.g., data bits DATA.18, DATA.19, . . . DATA.35) areavailable at the top of the memory array 102.

Each of the multiplexers 256, 257 is arranged to select all of itsrespective eighteen bits of data entirely from either the left bank 251or the right bank 253. Consequently, all 36 bits of data are availablefrom either the left bank 251 or the right bank 253. Because a separaterow decoder is associated with each bank, an active cycle may beperformed and yet only activate one bank of the memory array 102. Inother words, a selected word line may be driven high in one of the banksas part of performing an active cycle, while the other bank remainsinactive with no word line driven. Alternatively, because each of therow decoders 252, 254 has a row address bus independent of the other(not yet shown in FIG. 5), each row decoder may be supplied with adifferent row address. Thus, the two banks may both proceed withindependent active cycles. For example, a read cycle may proceed in theleft bank 251 in response to an externally requested cycle usingexternally-supplied address information, while an internal refresh cycleis performed in the right bank 253 using internally generated addressinformation.

FIG. 6 illustrates an embodiment of the multiplexer 256. An 18-bitbi-directional global data bus GDB.0, GDB.1, . . . GDB.17 is arranged tohorizontally run substantially across the width of the memory array 102(i.e., across the bottom of both the left array bank 251 and the rightarray bank 253). To save layout area, each bit of this global data busmay be implemented as a single-ended bus (each bit of data beingconveyed on a single wire) with a “rail-to-rail” data signal driventhereupon. To save power consumption, there are no load devices on theglobal data bus: each bus line is briefly either driven high to VDD orlow to VSS and then left dynamically floating and available to be sensedby a receiving circuit. To save further power, no reset or equilibrateis used on the global data bus lines. Rather, each bit of the globaldata bus is driven to the next data state or left at the previous datastate as required. A small, easily over-powered latch may be added toeach bit of the global data bus to maintain the state indefinitely untildriven to the other state.

Each of the 36 even-numbered global I/O lines (GI/O.0, GI/O.2, . . .GI/O.70) from the left array bank 251 is coupled to a respective globalI/O block (260.0, 260.2, . . . 260.70). The left-most 18 of these globalI/O blocks (260.0, 260.2, . . . 260.34) are connected respectively to acorresponding bit of the global data bus GDB.0, GDB.1, . . . GDB.17. Theright-most 18 of these global I/O blocks (260.36, 260.38, . . . 260.70)are also connected respectively to the corresponding bit of the globaldata bus (GDB.0, GDB.1, . . . GDB.17). Similarly, each of the 36even-numbered global I/O lines (GI/O.72, GI/O.74 (not shown), . . .GI/O.142) from the right array blank 253 is coupled to a respectiveglobal I/O block (260.72, 260.74 (not shown), . . . 260.142). Theleft-most 18 of these global I/O blocks (260.72, 260.74, . . . 260.106(not shown)) are connected respectively to a corresponding bit of theglobal data bus GDB.0, GDB.1, . . . GDB.17. The right-most 18 of theseglobal I/O blocks (260.108 (not shown), 260.110 (not shown), . . .260.142) are also connected respectively to the corresponding bit of theglobal data bus (GDB.0, GDB.1, . . . . GDB.17). Each of the global databus lines (GDB.0, GDB.1, . . . GDB.17) is connected to a correspondingdata I/O circuit (DATA.0, DATA.1, . . . DATA.17), which are each in turnconnected to a corresponding data I/O pad (PAD.0, PAD.1, . . . PAD.17).

The organization of the multiplexer 256 described above “spreads out” orspatially interdigitates 18 different 4:1 multiplexers across the bottomof the array 102. For example, the four different global I/O lines thatpotentially are connectable to bit 1 of the global data bus (i.e.,GDB.1) are GI/O.2, GI/O.38, GI/O.74 (not shown), and GI/O.110 (notshown). Such a spatial distribution requires, of course, each of theglobal data bus lines to traverse a greater length across the chip thanif the set of four global I/O lines connectable to a given global databus line were more localized. Nonetheless, at least two key advantagesresult from this arrangement shown in FIG. 6. First, each bit of the 18bits of data may be read from or stored into either the left array bank251 or the right array bank 253. This is one of the necessaryarchitectural requirements for a full 36-bit read or write cycle whichactivates only the left array bank or the right array bank, but notnecessarily both array banks. Secondly, this multiplexer arrangementensures that the global I/O lines corresponding to each 9-bit bytewithin the 36-bit data word are physically adjacent to each other. Forexample, the 9-bit byte conveyed on data I/O pads PAD.0, PAD.1, . . .PAD.8 is respectively coupled (for a given address) to the nineeven-numbered global I/O lines GI/O.0, GI/O.2, . . . GI/O.16, whichcommunicate with nine laterally adjacent read amplifier and write blocks202 within the selected hole of the memory array. As is described ingreater detail below, this adjacency may facilitate implementation of abyte write capability where one or more 9-bit bytes within a 36-bit dataword are individually selectable to be written, while leaving theremaining bits within the 36-bit data word undisturbed.

Having introduced at the block diagram level the organizational detailsof the local and global I/O lines, several individual circuits are nowdescribed. Referring again to FIG. 3, a bit line sense amplifier wasdescribed which is selectable to service two pairs of bit lines: oneabove and the other below the bit line sense amplifier. FIG. 4illustrates several of these bit line sense amplifiers, each of which isconnectable to a respective bit line pair within array block ARRAY.2,and connectable to a second respective bit line pair within either arrayblock ARRAY.1 or ARRAY.3. The portion of circuitry indicated in FIG. 4by a dashed block labeled 250 is illustrated by the schematic shown inFIG. 7. This detailed bit line sense amplifier schematic issubstantially identical to that described in FIG. 3 and indeed, whereidentical, uses the same element numbering. But FIG. 7 shows a fewadditional details and shows control signals which were omitted fromFIG. 4 in order to more easily illustrate the organization of local I/Olines and global I/O lines.

Referring now to FIG. 7, the earlier-described sense amplifier 142includes a pair of cross-coupled N-channel transistors 273, 274 whosecommon source terminals are driven toward ground by N-channel transistor275 when enabled by a high voltage on the strobe enable signal SEcoupled to its gate terminal. This common source node, NBIAS, is commonto many adjacent bit line sense amplifiers. Consequently, NMOStransistor 275 is actually a large distributed transistor pulling nodeNBIAS low for many bit line sense amplifiers.

The sense amplifier 142 further includes a pair of cross-coupledP-channel transistors 271, 272 whose common source terminals are driventoward VDD by P-channel transistor 270 when enabled by a low voltage onthe complementary strobe enable signal SEB (i.e., “strobe enable bar”)coupled to its gate terminal. This common source node, PBIAS, is commonto many adjacent bit line sense amplifiers. Consequently, PMOStransistor 270 is actually a large distributed transistor pulling nodePBIAS high for many bit line sense amplifiers.

The internal sense amplifier nodes SA, SAB are equilibrated to eachother by N-channel transistor 182 which is gated by the sense amplifierequilibrate signal SAEQ_LEVEL which remains high until the next activecycle begins in which sense amplifier 142 is selected. A common bit lineequilibrate node VBLEQ is equilibrated to the internal sense amplifiernodes SA, SAB by transistors 154, 158 which are gated by a self-timedpulsed sense amplifier equilibrate signal SAEQ_PULSE, to establish thebit line equilibration voltage of the collective high capacitance of allthe bit line pairs onto the relatively low capacitance of the common bitline equilibrate node VBLEQ, as described earlier.

The cross-coupled CMOS sense amplifier 142 is multiplexed to senseeither a bit line pair within the array block located above the senseamplifier 142, or to sense a bit line pair within the array blocklocated below the sense amplifier 142. To select the upper bit line pairBLU(i), BLBU(i), the upper array select signal ASU is left logicallyhigh and driven to VPP while the lower array select signal ASD, isquickly brought low. Array select transistors 160, 162 remain on andcouple the upper bit line pair BLU(i), BLBU(i) to respective senseamplifier nodes SA, SAB while array select transistors 164, 166 areturned off to isolate the sense amplifier nodes SA, SAB from the lowerbit line pair BLD(i), BLBD(i). Conversely, to select the lower bit linepair BLD(i), BLBD(i), the lower array select signal ASD is leftlogically high and driven to VPP while the upper array select signal ASUis quickly brought low to isolate the unselected bit line pair withinthe upper array block from the sense amplifier internal nodes SA, SAB.Since these array select signals which remain logically high (to gatethe selected bit lines to the respective sense amplifier) are preferably“boosted” to a voltage above the VDD level, the array select transistors160, 162, 164, 166 are preferably fabricated using a high voltagetransistor structure if one is available in the semiconductor processbeing used (and are so indicated by a “*” in the figure alongside eachtransistor).

The complementary bit lines BLD(i), BLBD(i) are equilibrated to eachother by equilibration transistor 156 which is gated by a self-timedpulsed equilibrate signal BLEQD and which transistor is located at the“near end” of the bit line pair (next to the array select transistors164, 166). Because the sense amplifier 142 restores one bit line to alow voltage level (i.e., ground) and the other bit line to a highvoltage level (e.g., 2.0 volts), the bit lines when subsequentlyequilibrated establish a bit line equilibration voltage that isapproximately one-half of the high write (i.e., restore) level. Each bitline pair BLD(i), BLBD(i) is also equilibrated at its “far end” (i.e.,the end most distant from its sense amplifier) by an equilibratetransistor gated by a decoded self-timed pulsed equilibrate signal BLEQUrunning through the next row of sense amplifiers (not shown in FIG. 7).The pulsed equilibrate signals at each end of a given pair of bit linesare active (i.e., “pulsed”) at the end of an active cycle only for theselected array block just completing its active cycle. The analogouspulsed equilibrate signals in the non-selected array blocks remaininactive (at VSS) without pulsing.

The other pair of complementary bit lines BLD(i+1), BLBD(i+1) which arelaid out in the pitch of this sense amplifier block 250, and which areserved by the sense amplifier in the next row of sense amplifiers below,are equilibrated to each other at their “far end” by equilibrationtransistor 277 which is also gated by the self-timed pulsed equilibratesignal BLEQD. The upper bit line pair BLU(i), BLBU(i) is similarlyequilibrated by equilibration transistor 170 which is gated by aself-timed pulsed equilibrate signal BLEQU and which transistor ispreferably located at the near end of the upper bit line pair BLU(i),BLBU(i) (next to the array select transistors 160, 162). The upper bitline pair BLU(i), BLBU(i) is also equilibrated at its far end by anequilibrate transistor (not shown) analogous to transistor 168 (see FIG.3) which is also gated by a decoded equilibrate signal having the sametiming as pulsed equilibrate signal BLEQU. Likewise, the other pair ofcomplementary bit lines BLU(i+1), BLBD(i+1) which are laid out in thepitch of this sense amplifier block 250, and which are served by thesense amplifier in the next row of sense amplifiers above, areequilibrated to each other at their “far end” by equilibrationtransistor 276 which is also gated by the equilibrate signal BLEQU.

The column select transistors 176, 178 are as previously describedrelative to FIG. 3. A single pair of column select transistors is laidout within each sense amplifier block 250, as shown, whose gates may beconnected to any one of the eight column select lines runninghorizontally through the sense amplifier layout cell. For example, eachof eight adjacent sense amplifier blocks 250 would be respectivelyconnected to a corresponding one of the eight column select lines (showndescriptively in FIG. 7 as COLSEL.X).

Referring briefly back to FIG. 4, sixteen sense amplifiers (such as thesense amplifier 250 described above) are coupled to and serviced by asingle local I/O-to-global I/O interface block 202, which provides theinterface to the associated global input lines GIN, GINB and globaloutput lines GOUT, GOUTB. One embodiment of a suitable localI/O-to-global I/O interface block 202 is illustrated in FIG. 18. Forread operations, the basic functions of this interface block 202 are toselect one of the two pairs of local I/O lines (e.g., either the leftlocal I/O pair LIO_L, LIOB_L, or the right local I/O pair LIO_R,LIOB_R), amplify the signal on the selected local I/O pair (which arenominally biased at a common mode voltage of approximately 1.0 volts),and drive the amplified signal onto the associated global output linesGOUT, GOUTB (which are nominally biased at a common mode voltage nearVDD). For write operation, the basic function of this interface block202 is to steer the driven signal on the associated global input linesGIN, GINB (which is nominally a 200 mV differential signal at a commonmode voltage of approximately 1.0 volts) onto the selected (left orright) local I/O pair (which are also nominally biased at the samecommon mode voltage).

A more complete description of the write operation of block 202 nowfollows. Two pairs of equilibrate transistors 185L, 186L and 185R, 186Rare used to equilibrate the two respective pairs of local I/O linesLIO_L, LIOB_L and LIO_R, LIOB_R to the bit line equilibrate potentialVBLEQ. These transistors are analogous to the single pair of equilibratetransistors 185, 186 first described in FIG. 3 for its single pair oflocal I/O lines. The sense amplifier equilibrate signal SAEQ_LEVEL isused for equilibrating the local I/Os and maintaining theirequilibration between active cycles.

During a write operation, the signal previously driven onto the globalinput lines GIN, GINB is steered by either transistors 180L, 181L ontothe left local I/O pair LIO_L, LIOB_L, or by transistors 180R, 181R ontothe right local I/O pair LIO_R, LIOB_R. Both steering pairs remaininactive if the particular global input line corresponds to a bit withina byte of the external data word which is disabled for writing.Otherwise, only one such steering pair is active at any one time. Forexample, to write to the left local I/O pair, a decoded write signalWRITE_L is driven high slightly before the column select lines aredecoded and driven so that node 312 is asserted high approximately atthe same time as the column select line is asserted (i.e., well beforebit line sensing). If the particular global input line corresponds to abit within a byte of the external data word which is enabled forwriting, then the byte write enable signal BYTE will have already beendriven high (by the write queue, as is described in greater detailherebelow). As a result, control node 312 is driven high by the actionof NAND gate 282 and inverter 284, and transistors 180L, 181L are turnedon to drive the signal present on the global input lines GIN, GINB ontothe left pair of local I/O lines LIO_L, LIOB_L. The asserted columnselect signal further steers this 200 mV differential write data signalonto the selected bit line sense amplifier internal nodes beforesensing, and the asserted column select signal then turns off as bitline sensing begins. To write to the right local I/O pair, a decodedwrite signal WRITE_R is driven high instead, along with the byte writeenable signal BYTE. As a result, control node 313 is driven high by theaction of NAND gate 283 and inverter 285, and transistors 180R, 181R areturned on to drive the signal present on the global input lines GIN,GINB onto the right pair of local I/O lines LIO_R, LIOB_R.

For read operations, one of the two pairs of local I/O lines is selectedand its signal is amplified by a two-stage amplifier 174 and driven ontothe associated global output lines GOUT, GOUTB. Rather than using passtransistors as in the write case, the left/right selection for read isaccomplished using a first stage amplifier having two differential inputtransistor pairs which are independently selectable by either a readleft signal READ_L or a read right signal READ_R. Between active cycles,the output nodes 286, 287 of the first amplifier stage are precharged toVDD by P-channel transistors 288, 289. These precharge transistors donot need to be gated (i.e., they are always “on”) because most suchamplifiers 174′ are inactive and no current flows therethrough. In therelatively few such amplifiers 174′ which are selected (i.e., active)for a given active cycle, the current flows through only one of theytransistors 288, 289 as a differential signal is developed on the firststage output nodes 286, 287, and the current flows for only a relativelyshort portion of the active cycle. Consequently, the extra complexityand power requirements of a decoded precharge signal is not necessary.Between active cycles, the decoded read signals READ_L, READ_R, and theleft/right non-decoded control signal READ are all low (inactive).Consequently, the output of inverter 304 is high, and transistor 303holds node 296 at ground. Similarly, the output of inverter 300, 301(i.e., node 314) is also high, and a VDD potential is developed acrossP-channel transistor 299, which is connected to function as a capacitor.In the second amplifier stage, equilibration transistor 305 is on toequilibrate the two cross-coupled nodes (the respective drains oftransistors 308, 309) to a threshold below VDD, and to pull thecommon-source node (drain of transistor 310) to a voltage of VDD lesstwo thresholds.

During an active read cycle, the READ signal (for the selected senseamplifiers) is driven high and the left/right decoded read signal READ_Lor READ_R is also driven high, both at about the same time as thedecoded one of the column select lines is driven high (e.g., well beforebit line sensing). As an example, assume one of the eight senseamplifiers connected to the left local I/O line pair is to be read. Whenthe READ signal is driven high, the precharge transistor 303 in thefirst stage turns off, as does the equilibrate transistor 305 in thesecond stage. Additionally, the inverter 300, 301 switches and begins todrive its output node 314 low. Transistor 302 receives on its gateterminal a DC bias voltage V15V having, for example, a nominal value of1.5 volts. Consequently, an essentially constant current flows throughthis transistor 302 as long as it remains saturated (as long as itsdrain voltage exceeds its gate-to-source voltage less its thresholdvoltage). The inverter 300, 301, the constant current transistor 302,and the capacitor (transistor) 299 are employed to provide at node 296what appears to be a constant current source connected to a negativevoltage. (which, as described below, provides a “tail” current for theselected differential transistor pair).

Because the decoded signal READ_L is also driven high, transistor 298 isconductive and the differential pair formed by transistors 293 and 290is enabled to amplify the signal present between their respective gates,namely LIO_L and LIOB_L, and to develop an output signal on the firststage output nodes 286, 287. The other differential pair formed bytransistors 291, 292 remains off since READ_R remains low and transistor297 remains off. To read the right pair of local I/O lines LIO_R,LIOB_R, the decoded signal READ_R is driven high while READ_L remainslow. The “right” differential pair formed by transistors 291, 292 isenabled through transistor 297, and the “left” differential pair formedby transistors 290, 293 remains disabled.

As is well known in the art, a basic differential amplifierconfiguration is advantageously implemented using a constant currentsource in the “tail” of the differential pair. Frequently, such aconstant current source is approximated by a single N-channel transistorwith a DC bias voltage on its gate having a value which is less than thenominal common mode voltage of the two input signals connected to thegates of the differential pair. Such a configuration assumes that thecurrent tail transistor remains saturated, which requires its drainvoltage to exceed its gate voltage less a threshold. But in atraditional differential pair configuration, the drain voltage of thecurrent tail transistor (assuming it's the same node as thecommon-source node of the differential pair) must be lower in voltagethan the higher of the two input signals less a threshold voltage forany current to flow through either of the differential pair transistors.As a result, it is exceedingly difficult to use a traditional NMOSdifferential pair, having manufacturable threshold voltages, to sense asignal having a very low common-mode voltage which approaches thethreshold voltage. In the embodiment shown, the nominal equilibrationvoltage of the local I/O lines may be as low as 0.75 volts (althoughpreferably it is around 1.0 volts, as described below). With N-channeltransistor threshold voltages also in the range of 0.75, such asamplifier might not have any conductive paths, and therefore obviouslycould not be guaranteed to properly function.

To afford the capability of sensing signals having such a low commonmode voltage, the inverter 300, 301, the constant current transistor302, and the capacitor (transistor) 299 are employed to provide at node296 what appears to be a constant current source connected to a negativevoltage. When the READ signal is driven high, node 314 falls in voltagefrom VDD toward ground, and node 296 falls from ground to a negativevoltage (since it was precharged to ground, but since transistor 303 isnow off). As soon as the voltage on node 296 and on either node 294 or295 (whichever is selected) are low enough for one side of the selecteddifferential pair to start conducting, the voltage at node 296 issubstantially clamped at that voltage. Since transistor 302 (and 301)provide a constant current to discharge node 314, the voltage of node314 falls with a linear ramp. Since node 296 is essentially clamped to aconstant voltage, the change in voltage across capacitor 299 istherefore linear with time. Consequently, the displacement currentthrough capacitor 299 (i.e., the saturated current of transistor 302)drives 296 negatively by whatever voltage is required (with certainlimitations, of course) for the sum of the currents of the differentialpair to equal the current through transistor 302. The actual voltageresulting on node 296 depends on the transistor threshold voltage andthe particular voltages present on the gates of the differential pair.For some operating conditions, this voltage may be above ground ratherthan below ground, but the “tail” current of the differential pairnonetheless remains equal to the saturated current of transistor 302.

The first stage amplifier, therefore, accomplishes the difficult task ofsensing and amplifying a signal having a low common mode voltage, anddeveloping an amplified signal on output nodes 286, 287 which are biasednear VDD. Furthermore, it accomplishes this using relatively fast NMOStransistors rather than using slower PMOS transistors whose biasingwould have been more straightforward. The second stage amplifier(transistors 305, 306, 307, 308, 309, 310, and 311) further amplifiesthe signal on its inputs (nodes 286, 287) and drives the amplifiedsignal onto the global output lines GOUT, GOUTB. The amplifier isequilibrated when READ is low by transistor 305, which equilibrates thetwo internal circuit nodes 315, 316 (and which nodes are precharged to athreshold below VDD by transistors 306, 307 since the global outputlines GOUT, GOUTB and nodes 286, 287 are all precharged to VDD). WhenREAD is driven high, the equilibrate transistor 305 turns off and thecurrent source transistor 311 is coupled through transistor 310 toprovide a constant current to the non-latching cross-coupled transistors308, 309. As configured here, the two input nodes 286, 287 areprecharged to VDD, and one of them falls slightly in voltage when signalis developed by the first stage amplifier. The other input node remainsat VDD. As the amplifier starts to sense, the source of the NMOStransistor having the lesser input gate voltage is brought low, whilesource of the NMOS transistor having the higher input gate voltage(i.e., the side whose input gate remains at VDD) stays high. Because ofthe voltage amplification of this second stage amplifier, the internalnode 315 or 316 which falls in response to the falling input voltage(node 286 or 287) actually falls quite a bit further than does the inputnode. Due to large gate-to-source capacitance of the input transistors(especially because they are biased in saturation) there is a largenegative coupling from the transistor source back into the input gatenode. For example, if node 287 is falling from VDD to 200 mV below VDD,node 316 may fall, for example, by 800 mV, or 4 times as much as theinput falls. The charge removed from the input node provides a negativeinput capacitance of the second stage amplifier onto the output nodes ofthe first stage amplifier, thus speeding up the response of the firststage amplifier. The differential voltage between nodes 286 and 287provides a much larger differential voltage between nodes 315 and 316,which causes differential currents through transistors 309 and 308which, in turn, provide a differential current pulling either GOUTB orGOUT low. These and other features and advantages of this second stageamplifier are described more fully in co-pending application Ser. No.09/223,265, filed on Dec. 30, 1998, naming Robert J. Proebsting asinventor and entitled “Differential Sense Amplifier Circuit,” thedisclosure of which is incorporated herein by reference in its entirety.

Referring now to FIG. 9, a global I/O interface block 260.X is shownwhich was introduced in FIG. 6 and which, when reading, senses a smallsignal on a pair of global output lines GOUT, GOUTB and drives asingle-ended global data bus accordingly with a rail-to-rail signalswing, and which, when writing, receives a data bit from the global databus and drives a small differential signal onto a pair of global inputlines GIN, GINB which were previously equilibrated to the global inputline equilibrate voltage VGINEQ.

Taking the more straightforward read path first, the pair of globaloutput lines GOUT, GOUTB are precharged to VDD by P-channel loadtransistors 191, 192, first introduced in FIG. 3, which also serve asactive load devices. A latching sense amplifier 193 includes a full CMOScross-coupled latch (transistors 323, 324, 325, and 326) whosecross-coupled nodes 329, 330 (which are also the output nodes) areprecharged to VDD by transistors 321, 322 when a LATCH signal is low(inactive), and which cross-coupled latch is caused to latch a “0” or a“1” (i.e., “steered”) by transistors 327 and 328 according to thedifferential signal on the global output lines GOUT, GOUTB when LATCHgoes high (active). Since even the low-going global output line (eitherGOUT or GOUTB) never drops much below VDD (and the other global line, asdescribed above, stays substantially at VDD) the cross-coupled latchpulls one of its outputs (node 329 or 330) all the way to ground, whilethe other output (node 330 or 329) is held at VDD. As long as acomplementary read strobe signal RBSTB remains inactive (high), theoutputs of both NOR gates 332 and 333 are driven low, and push-pulldriver transistors 335 and 336 are both held off. When the read strobesignal RBSTB is driven active (low) after the LATCH signal is drivenhigh, the global data bus GDB is driven according to which cross-coupledoutput node 329, 330 is low. If node 330 is latched low, the gate ofNMOS transistor 336 is driven high through NOR gate 333. Thus,transistor 333 drives the global data bus GDB to ground, while the gateof P-channel transistor 335 is held high by inverter 334. On the otherhand, if node 329 is latched low, the gate of transistor 336 is held lowby NOR gate 333 while the gate of P-channel transistor 335 is driven lowby NOR gate 332 and inverter 334, thus driving the global data bus toVDD. In either case, the complementary read strobe signal RBSTB isactive for a long enough time to drive a full rail-to-rail signal on thesingle-ended global data bus GDB (e.g., drive the voltage fully toeither a VDD or ground level). There is no precharge circuit for theglobal data bus: it is driven with the next data on the next cycle. Iftwo sequential cycles drive the same data onto the global data bus, thenthe voltage of the global data bus doesn't change. By not equilibratingor precharging this bus, which likely traverses across a significantportion of the width of the memory array, significant power is savedduring an active cycle.

In a write operation, a data bit is placed onto the global data bus atthe appropriate time by the write queue, which is described furtherbelow. For the description of the global input line driver shown here inFIG. 9, assume the global data bus is already driven either to VDD or toground in accordance with the data to be written onto the global inputline GIN, GINB (and ultimately into the selected memory cell). Inverters337 and 338 buffer and invert the data received on the global data busGDB to create complementary local nodes corresponding to the data on theglobal data bus GDB. When the write bus strobe signal WBST momentarilygoes high (active), the complementary data is loaded via transistors339, 340 into the latch formed by inverters 341 and 342 by over-poweringthe latch. Thus, complementary data is available, after the pulse ofstrobe signal WSTB, on nodes 343 and 344 which are respectivelyconnected to NAND gates 345, 346.

The global input line driver circuit 190 receives the complementary datasignals conveyed on nodes 343, 344, as well various control and timingsignals and drives the global input lines GIN, GINB with a constantcurrent pulse of a controlled magnitude for a timed duration to providea reasonably predictable change in charge on each of the global inputlines. A positive charge added to one of the global input lines GIN orGINB and a negative charge on the other creates a predictable smalldifferential signal between GIN and GINB. In FIG. 9, two pairs of globalinput lines are illustrated, namely GIN0, GIN0B and GIN1, GIN1B. Asdescribed earlier, the pairs of global input lines are interdigitated ineach bank of the memory array, with half of the pairs exiting the memorybank toward the I/O section above the bank, and the other half exitingthe memory bank toward the I/O section below the bank. As shown in FIG.9, the left-most global input pair GIN0, GIN0B represent a pair whichexits at the opposite end of the memory bank. The transistors 187′,188′, and 189′ form a “far end” equilibration circuit for this pair ofglobal input lines, which are driven by a driver circuit (e.g., likedriver circuit 190) and also equilibrated at the other end of the lines(i.e., the “near end”) on the opposite side of the memory bank usinganother circuit as shown in FIG. 9. The transistors 187, 188, and 189form a “near end” equilibration circuit for the second pair of globalinput lines, namely GIN1, GIN1B.

The equilibration signals GINEQ0 and GINEQ1 are driven high, if drivenat all for a particular active cycle, during the latter portion of bitline sensing to equilibrate the global input lines together. Theequilibration voltage for the global input lines is set by the commonglobal input line equilibration node VGINEQ, which is coupled to thememory cell plate voltage PLATE which itself is driven by asample-and-hold amplifier (e.g., see amplifier 196 in FIG. 3) to thesame voltage as the bit line equilibrate voltage VBLEQ. Consequently,the global input lines are equilibrated to very nearly to the samevoltage as are the bit lines (nominally about 1.0 volts). The commonglobal input line equilibration node VGINEQ is coupled (duringequilibration) to all pairs of global input lines receiving new writedata to also help ensure a uniform equilibration voltage to all suchpairs of global input lines.

The equilibrate signal GINEQ does not pulse for every active cycle, buttypically only after a write cycle, and only for those global inputlines receiving new write data as is described in greater detail below.To understand the operation of the driver circuit 190, assume that theglobal input pair GIN1, GIN1B has been equilibrated to a 1.0 volt level,and that the equilibration signal GINEQ1 is now low and transistors 187,188, and 189 consequently are each turned off. As described above, alsoassume that complementary data to be written is already available onnodes 343 and 344.

As mentioned above, the global input line driver circuit 190 drives theglobal input lines GIN1, GIN1B with a constant current pulse for a timedduration to create a predictable small voltage signal between GIN1 andGIN1B. In operation, this write data input signal is driven onto thepair of global input lines (frequently well before the actual writeoperation in which the data is actually written into the selected memorycell), and the global input lines are then allowed to float, in effectstoring dynamically on the pair of global input lines the next data bitto be written by the next write operation, even if several readoperations may occur before the next write operation. A more completedescription of the operation of the write queue, which makes the data tobe written available to the driver circuit 190 well before the actualwrite operation is typically performed internally in the memory array,follows below.

The control signal WGIN is a timing pulse that controls the length ofthe current pulse applied to the global input lines. When inactive, thecontrol signal WGIN is low, which holds the respective outputs 347, 348of NAND gates 345, 346 high. Consequently, the gates of P-channeltransistors 351, 352 are high and the gates of N-channel transistors349, 350 are low. Thus, all four transistors (349, 350, 351, and 352)connected to either of the global input lines GIN1, GIN1B are off, andthe driver circuit 190 presents a high impedance to the global inputlines GIN1, GIN1B and allows the global input lines to either float(which preserves the data signal previous developed thereon) or beequilibrated by both the “near-end” equilibration circuit (transistors187, 188, and 189) and by a “far-end” equilibration cilrcuit (not shownin FIG. 9, but analogous to transistors 187′, 188′, and 189′).

Each of the global input lines (e.g., GIN1, GIN1B) is allowed to floatuntil the corresponding internal write operation actually utilizes thedata signal thereon, and moreover until a subsequent external writecycle presents data to be written to a data bit which corresponds to theparticular global input line. To ensure that the dynamically storedsignals do not decay significantly from the global input lines shouldthe next external write cycle not be presented to the chip within areasonable time, the write signals thereon are refreshed periodically byequilibrating the global input lines and re-developing the writesignals. In the preferred embodiment, the internal row refresh counterwhich generates an output every 256 clock cycles is used to initiatesuch a global input line refresh, whether or not the particular memorybank is actually active with a refresh cycle, is active with anon-refresh cycle, or is even otherwise inactive. The global input linerefresh need not be synchronized to any row refresh activity because theglobal input line decay (if any) is substantially independent of whetherany memory cells need refreshing. Consequently, any other counter orother means of generating a suitable refresh interval may also be usedto initiate a refresh of the global input lines.

To write new data onto the global input line pair (or to refresh thedata already there), it is first equilibrated by application of theequilibration signal GiNEQx (i.e., GINEQ0 or GINEQ1), and then (againassuming that the new data to be written has already been strobed off ofthe global data bus GDB and is already present on complementary nodes343, 344) the timing pulse WGIN is applied to the driver circuit 190.For each particular write cycle, if the global input line paircorresponds to a bit within a byte which is byte-enabled for writing,then the local byte write enable signal LBW (received by the driver 190for such a global input line pair) is high. For any other byte,including other bytes not even within the presently addressed data word,the local byte write enable signal LBW remains low, and the timing pulseWGIN is ignored by NAND gates 345, 346. However, for a global input linewhich is byte-enabled for writing (i.e., in which LBW is also high),either NAND gate 345 or NAND gate 346 will progogate the timing pulsereceived on timing signal WGIN and effectuate the write to the globalinput line pair. The LBW signal is preferably decoded from externalcontrol signals and is set up with about the same timing as the datasignals. Both are preferably strobed by the write bus strobe signalWBST.

To better describe the operation of driver 190 in generating a smalldifferential signal on the global input line pair, assume that a “1” isto be driven onto the global input line pair GIN1, GIN1B. Accordingly,node 343 is low, and node 344 is high. Also assume, of course, that thelocal byte write enable signal LBW is active (high): otherwise, no writewould occur, as described above. When the high-going timing pulse WGINarrives, the output node 348 of NAND gate 346 switches low, since allthree of its inputs are now high. The low voltage on node 348 is appliedto the gate of P-channel transistor 351. The gates of transistors 355and 356 are both biased at a DC bias voltage WPBIAS, which is locally RCfiltered by N-channel transistor 358 (connected so as to largely act asa resistor) and transistor 359 (connected to function as a capacitor toVDD). In this configuration, transistors 355 and 356 each implement arather effective constant current source able to source a predictable(and fairly VDD independent) amount of current from the VDD supply intorespective nodes GIN1 and GIN1B when enabled by respective switchingtransistors 351 and 352. Assuming the data polarity of the presentexample, since node 348 is low, the series combination of current-sourcetransistor 355 and switch transistor 351 allows a predictable amount ofcurrent to flow from VDD into node GIN1 for the duration of the highpulse on the timing signal WGIN. Since transistor 349 is still off andno current is conducted from node GIN1, the integral of the current forthe duration of the pulse results in a predictable increase in charge ofnode GIN1. The voltage of the global input line GIN1 (i.e., node GIN1)increases by an amount equal to this change in charge divided by thetotal capacitance of the global input line GIN1. In the embodimentdescribed, the length of the timing pulse WGIN, and the width andgate-to-source voltage of the current source transistor 355 are suchthat the value of this change in voltage on the global input line GIN1is approximately +100 mV.

The value of the internally-generated bias voltage WPBIAS is preferablyset to achieve the greatest amount of bias (i.e., VGS) across each oftransistors 355 and 356 and yet still bias each transistor insaturation. Since the desired high level of the global input line ispreferably equal to the equilibration voltage (e.g., 1000 mV) plus a 100mV write signal (e.g., for a total voltage of 1.1 volts on a “writtenhigh” global input line), the value of the bias voltage WPBIAS ispreferably set to approximately the magnitude of one PMOS thresholdvoltage below this 1.1 volt desired high level. By maximizing themagnitude of the gate-to-source bias voltage presented to the currentsource transistors 355, 356, at least four advantages result: (1) for adesired amount of current, the width of each current source transistoris reduced (2) for a given width of the current source transistor, thewidth of the timing pulse WGIN which is necessary to achieve the desiredwrite signal on the global input lines is reduced; (3) the sensitivityof the current source value to internal chip noise is reduced; and (4)the effectiveness of the local RC decoupling circuit (transistors 358,359) in reducing drain-to-gate coupling is enhanced.

Looking at the lower half of the driver 190, the gates of transistors353 and 354 are both biased at the voltage WNBIAS, which is preferablyan internally-generated bias voltage relative to VSS (ground) having avalue of approximately (0.9+V_(TN)) volts above VSS, and which for thisembodiment is locally RC filtered by P-channel transistor 360 (connectedso as to largely act as a resistor) and N-channel transistor 361(connected to function as a capacitor to VSS). In this configuration,transistors 353 and 354 each implement a rather effective constantcurrent source able to sink a predictable (and internal chip noiseinsensitive and fairly non-power supply dependent) amount of currentfrom the respective nodes GIN1 and GIN1B into the ground node whenenabled by respective switching transistors 349 and 350. Again assumingthe data polarity of the present example, since node 348 is low, thegate of transistor 350 is high and the series combination ofcurrent-source transistor 354 and switch transistor 350 allows apredictable amount of current to flow from node GIN1B to ground for theduration of the high pulse on the timing signal WGIN. Since transistor352 is still off (node 347 is high for this data polarity) and nocurrent is conducted from VDD into node GIN1B, there results apredictable decrease in voltage of GIN1B. In the preferred embodimentdescribed, the length of the timing pulse WGIN, and the width andgate-to-source voltage of the current source transistor 354 are suchthat the value of this decrease in voltage on GIN1B is alsoapproximately 100 mV.

The value of the internally-generated bias voltage WNBIAS is preferablyset to achieve the greatest amount of bias (i.e., VGS) across each oftransistors 353 and 354 and yet still bias each transistor insaturation. In the preferred embodiment, since the desired written lowlevel of the global input line is preferably equal to the equilibrationvoltage (e.g., 1000 mV) minus a 100 mV write signal (e.g., for a totalvoltage of 0.9 volts on a “written low” global input line), the value ofthe bias voltage WNBIAS is preferably set to approximately (0.9+V_(TN))volts above VSS. By maximizing the bias voltage presented to the currentsource transistors 351, 352, the same advantages result as describedabove for the pull-up current sources. Note that the duration of thetiming signal WGIN is equal to both the duration of the positive currentsource pulling GIN1 up, and the duration of the negative current sourcepulling GIN1B down. Since it is desired to increase the voltage of GIN1Bby the same amount that the voltage of GIN1 is decreased, the magnitudeof the positive and negative current sources are preferably equal.

At the conclusion of the timing pulse WGIN, a 200 mV differentialvoltage exists between the two lines of a global input line pair, havinga polarity in accordance with the data bit to be written on the nextwrite operation. (While the operation of driver circuit 190 wasdescribed above in the context of a particular exemplary data polarity,the driver 190 is a symmetrical circuit and its operation with theopposite data polarity need not be described in detail.) The two globalinput lines GIN1, GIN1B are left floating dynamically until the nextwrite operation is actually carried out, which may be some time in thefuture (e.g., several active cycles later). Assuming a preferred globalinput line equilibration voltage of 1.0 volts, the preferred “high”voltage on a global input line is therefore 1.1 volts, and the preferred“low” voltage on a global input line is therefore 0.9 volts. Thesepreferred voltages are chosen because they are sufficient to cause a bitline sense amplifier to latch in accordance with the write data, yet aresmall enough to not disturb the sensing of neighboring bit lines.

It should also be emphasized that, because of the write cycle “merging”capability of the preferred embodiment (which is described in greaterdetail below), a data strobe signal WBST may be provided to latch therespective data (from the global data bus GDB) corresponding to a firstexternal write cycle into the respective driver circuits 190 for a firstdecoded group of global input lines, followed by a pulse of the timingsignal WGIN applied to the respective driver circuits 190 for the firstgroup of global input lines to drive the respective latched data ontothe respective global input lines. Then, in a subsequent cycle, anotherWBST pulse is provided to latch the respective data corresponding to asecond external write cycle into a second set of respective drivercircuits 190 for a second decoded group of global input lines, which isthen followed by a pulse of the timing signal WGIN applied to therespective driver circuits 190 for this second group of global inputlines to drive the respective latched data onto the respective globalinput lines. A single internal write operation then simultaneouslywrites the data for both groups into their respective memory cells(decoding and driving a word line, steering the two groups of selectedsense amplifiers with the two groups of write data already present onthe global input lines, restoring the data as latched in the senseamplifiers into the selected memory cells, etc.).

Alternatively, a data strobe signal WBST may be provided to latch therespective data (from the global data bus GDB) corresponding to a firstexternal write cycle into the respective driver circuits 190 for a firstdecoded group of global input lines, followed by another WBST pulse tolatch the respective data corresponding to a second external write cycleinto a second set of respective driver circuits 190 for a second decodedgroup of global input lines, then followed by a pulse of the timingsignal WGIN applied to the respective driver circuits 190 for bothgroups of global input lines to drive the respective latched data ontothe respective global input lines, and only then followed by a singleinternal write operation to write the two groups of write data alreadypresent on the global input lines.

Such write cycle merging may be performed when consecutive write cyclesaddress portions of the same 72-bit double word (i.e., when therespective row and column address portions of consecutive write cyclesdecode to the same column of the same word line (i.e., row) within thesame array block within the same memory bank). In other words, writecycle merging may occur when consecutive write addresses differ only inthe least significant address bit (the LSB) used by the multiplexer 109shown in FIG. 1. This is particularly attractive when accessing thememory using sequential addresses, as would frequently occur during aburst mode access or when accessing a contiguous block of data, such asa cache line fill operation for a processor. Moreover, there is noreason to limit cycle merging at just two consecutive cycles. As anadditional example, four consecutive external write cycles, eachaddressing a different 9-bit byte of the same addressed 36-bit word,followed by another four consecutive external write cycles, eachaddressing a different 9-bit byte of its addressed 36-bit word, willresult in only one internal write operation so long as the two writeaddresses differ only in the least significant column address bit. Thepower savings from such a “merging” of write cycles is extremelysignificant. Moreover, such merging keeps the ultimately selected memorybank inactive during the “merged” cycle, which allows a hidden refreshcycle to occur in the selected memory bank during the “merged” cycle.Additional details and advantages of write cycle merging and of thewrite queue are described further below.

The array architecture described above, including the configuration andarrangement of the array blocks, shared sense amplifiers, local I/Olines serving several sense amplifiers from a global input and output“quad” of lines (an input pair GIN, GINB and an output pair GOUT,GOUTB), and the individual supporting circuits implementing thisarchitecture which are also described above provides an attractivecompromise between performance and memory array efficiency (i.e., thepercentage of the overall memory chip which is actually memory cellsrather than the necessary support circuitry such as sense amplifiers,decoders, input/output circuits, control circuits, other peripheralcircuits, etc.)

In another embodiment illustrated beginning in FIG. 10, a higherperformance is achieved, albeit at the inevitable expense in arrayefficiency (and therefore, die size), by replacing the pairs of localI/O lines LIO, LIOB with a dedicated pair of local output lines and adedicated pair of local input lines, and by essentially implementing aportion of the shared first stage amplifier within amplifier 174′ (FIG.8) into each and every bit line sense amplifier.

Referring specifically to FIG. 10, a first stage amplifier portion 371is implemented within each and every bit line sense amplifier (e.g.,within bit line sense amplifier 250 shown in FIG. 4 and again in greaterdetail in FIG. 7) while the remaining portions of the first stageamplifier, the left/right multiplexer, and the second stage amplifier(all collectively indicated as circuit block 372) are implemented, asbefore, within the local I/O block 202 (see FIG. 4). In particular, thebi-directional local I/O section 278 of every bit line sense amplifier(shown in FIG. 7) is replaced by the first stage amplifier portion 371for reading, and by additional write circuitry described below.

In the first stage amplifier portion 371, the internal sense amplifiernodes SA, SAB are directly connected to the gates of a differential pair(transistors 373 and 374, respectively) rather than, in the earlierembodiment, coupled first through column select transistors onto a localI/O line pair, and then to the differential pair. By being directlyconnected to the relatively low-capacitance of the transistor gateterminals, the capacitive loading on the internal sense amplifier nodesSA, SAB is greatly reduced since it no longer includes the capacitanceof the local I/O lines (and the DC loading, of course, is virtuallyzero). The selection of a particular first stage amplifier isaccomplished by a first switch transistor in the current tail within thefirst stage amplifier portion 371 (e.g., transistor 375) which is drivenby one of the column decode signals, and further by a second switchtransistor in the circuit block 372 (e.g., transistor 377) which isdriven by a left/right read select signal (e.g., RD_L). The magnitude ofthe current tail current through the first stage differentialamplifiers, as before, is largely determined by a current sourcetransistor with a DC bias on its gate, in this embodiment shown as anominally 1.5 volt signal V15V applied to both transistors 378 and 380within circuit block 372. As before, if the voltages to be sensed aretoo low relative to the threshold voltages of the differential pair oftransistors, node 376 (or node 389) can be driven below ground, forexample, by circuitry included in amplifier 174′ shown in FIG. 8.

The left-most eight first stage amplifiers are coupled to a shared localoutput pair LOUT_L, LOUTB_L which physically runs through each of theeight sense amplifiers. A single pair of load “resistors” (e.g.,grounded-gate P-channel transistors 381, 382) is provided for the leftlocal output pair LOUT_L, LOUTB_L within the circuit block 372.Similarly, the right-most eight first stage amplifiers are coupled to ashared local output pair LOUT_R, LOUTB_R which physically runs througheach of the eight right-most sense amplifiers. A single pair of load“resistors” (e.g., transistors 383, 384) is provided for the right localoutput pair LOUT_R, LOUTB_R within the circuit block 372. A 2:1multiplexer formed by P-channel select transistors 385, 386, 387, and388 couples either the left or right pair of local output lines to acommon pair of nodes 286, 287, which are the input nodes to the second,stage amplifier (transistors 305, 306, 307, 308, 309, 310, and 311), asbefore.

Between active cycles, both read select signals RD_L and RD_R areinactive low, as are each of the column select signals (e.g., CS.1).Consequently, each of the first stage amplifiers 371 are off,intermediate nodes 376 and 389 are floating, and the local output linesLOUT_L, LOUTB_L, LOUT_R, and LOUTB_R are driven to VDD by loadtransistors 382, 381, 384, and 383, respectively. Because bothleft/right read select signals RD_L and RD_R are low, P-channeltransistors 385, 386, 387, and 388 are on, which ensures that the outputnodes of the multiplexer (i.e., the input nodes 286, 287 of the secondstage amplifier) are also brought fully to VDD.

In an active cycle, one of the two left/right read select lines RD_L orRD_R, one of the eight column select lines CS.X, and the READ signal areall driven high at or about the same time. For example, assume that thefirst stage amplifier portion 371 is to be read during an active cycle.The column select signal CS.1 is driven high at about the same time asthe left/right read select signal RD_L is driven high. The twomultiplexer transistors 387, 388 are turned off, isolating the rightlocal output pair LOUT_R, LOUTB_R from the common node pair 286, 287.Current flows through transistors 378, 377, and 375 to provide a tailcurrent for differential pair 373, 374, which develops an output voltageon left local output pair LOUT_L, LOUTB_L. This signal is communicatedthrough multiplexer transistors 386, 385 to the input nodes 286, 287 ofthe second stage amplifier, which functions as described above toamplify and drive the signal onto the global output pair GOUT, GOUTB.One of the eight right-most first stage amplifiers is also selected bythe same column select line (for example, CS.1 for this description),but transistor 379 is off because the left/right read select signal RD_Ris low. Therefore, no signal is developed on the right local output pairLOUT_R, LOUTB_R. The size of P-channel multiplexer transistors 385, 386,387, and 388 are advantageously kept small to take advantage of thenegative input capacitance of the second stage amplifier, describedearlier with respect to FIG. 8.

The corresponding circuitry, for this embodiment, for implementing awrite operation is illustrated in FIG. 11. A write block 401 is providedfor each bit line sense amplifier 250 of FIG. 7 (and implemented alongwith the first stage amplifier portion 371 of FIG. 10 described abovewithin each sense amplifier 250, in place of the I/O section 278). Whilehigher in performance (due to the much lower capacitance on the internalsense amplifier nodes), the area required to implement a sense amplifierincluding the first stage amplifier portion 371 and the write block 401(totaling 7 transistors) is much larger than a sense amplifier includingthe bi-directional I/O section 278 (as in FIG. 7) which totals just 2transistors. The global input pair GIN, GINB corresponding to the givengroup of bit lines, and which pair is preferably implemented in the toplayer of metal and runs vertically through the array bank (e.g., arraybank 251 shown in FIG. 5), is connected in each hole between arrayblocks to also run horizontally (in preferably a lower layer of metal)over a length equal to sixteen bit line sense amplifiers (eighttypically to the left of the vertical GIN, GINB pair, and eighttypically to the right). In the embodiment shown, the connection block402 (simply a via or contact between layers of metal) is associatedwithin the local I/O read amplifier and write block 202 (see FIG. 4),while the write block 401 is repeated for each of the sixteen bit linesense amplifiers 250 which share a given local I/O read amplifier andwrite block 202. When enabled, a given write block 401 couples the datainput signal conveyed on the global input lines GIN, GINB to thecorresponding internal sense amplifier nodes SA, SAB. As describedabove, this occurs before bit line sense amplifier sensing, which allowsthe sense amplifier itself to actually write the high or low level intothe selected memory cell.

A write block 401 is enabled by the coincident application of threelogical signals: the decoded one of the eight column select signals(e.g., CS.1) which are also used by the read circuitry, one of twoleft/right write select signals WR_L, WR_R, and a third signal, BW(“byte write”), which is enabled when the particular bit correspondingto the group of sixteen bit line sense amplifiers corresponds to anexternal byte which is enabled for writing. To more fully describe theoperation, assume a bit line pair associated with the write block(labeled as 401) is to be written with a logical “1”. In such a case,the global input lines GIN, GINB would already have been driven byoperation of the write queue to reflect the data to be written into theselected memory cell during the “next” write cycle, as was describedabove. In this example, the true global input line GIN would havealready been driven (for this embodiment) to a voltage equal to theequilibrate level (e.g., 1.0 volts) plus 100 mV, for a total of 1.1volts, while the complementary global input line GINB would have alreadybeen driven (for this embodiment) to a voltage equal to the equilibratelevel less 100 mV, for a total of 0.9 volts. The global input lines,after being driven to these voltages, remain dynamically floating afterthe write cycle occurs which actually uses the data until just beforethe next write cycle which uses the same global input lines, or untilthe global input lines are refreshed, whichever comes first. In eithercase, such global input lines are equilibrated and driven with thecorresponding write data signal to create the appropriate differentiallevels. Moreover, the four byte write control signals for the next writedata are driven high (to enable) or low (to disable) writing to the ninebits associated with the respective 9-bit byte. Consequently, the,corresponding byte write signal BW would already have been driven byoperation of the write queue in accordance with the address of theselected memory cell for the “next” write cycle (e.g., BW is drivenhigh, or remains high if already so, in any hole which is enabled forwrite and the particular byte is selected for writing).

During the active write operation, the byte write signal BW is alreadyvalid. The equivalent of the 2:1 multiplexer found in the read path isaccomplished in the write path by using the two left/right write controlsignals WR_L and WR_R. The left or right component of this informationis valid in advance from the write queue. Since the selected arrayselect and the selected hole(s) is also stored in the write queue andavailable in advance when a write cycle begins, this information isalready combined, waiting for the next write cycle. As soon as the nextwrite cycle is decoded and as early in the cycle as possible, either theWR_L or the W_R signal for the active hole(s) is enabled. These WR_L andWR_R signals are normally low (to prevent writing during a read cycle),and pulse high during an active write cycle when selected. As isdescribed below, there will normally be one selected hole for writingduring any one write cycle, either the hole above or the hole below theselected array block. In the case of a merged write, both the hole aboveand the hole below the selected array block are enabled for writing, andthe WR_L or WR_R signal and the selected column select signal for bothholes will be driven high. If a byte write signal BW for a particularglobal input line is high, then the selected column to the left or rightis going to be written by the next write operation, whenever it occurs,with write data corresponding to the global input line.

The eight left-most bit line sense amplifiers associated with thisglobal input pair GIN, GINB are coupled to receive the left writecontrol signal WR_L at the drain of transistor 403, while the eightright-most bit line sense amplifiers associated with this global inputpair GIN, GINB are coupled to receive the right write control signalWR_R at the drain of the analogous transistor. Looking at the writeblock 401 on the left side of FIG. 11, the gate of transistor 404 isdriven high when both the WR-L signal is driven high with the BW signalalready high. Then, after the gate of transistor 404 is driven high, thecolumn select signal CS.1 is driven high, and the voltage on the gate oftransistor 404 bootstraps up well above VDD by thle channel capacitanceof transistor 404 so that the gate node of transistors 405 and 406(i.e., node 407) is driven to follow the column select signal, withoutlosing a threshold voltage in going through pass transistor 404. Inparticular, the VDD-level voltage communicated onto node 407 is highenough to drive an internal sense amplifier node (e.g., SA) from anequilibrate level of 1.0 volts up to a pre-sense write level of 1.1volts as it drives the complement sense amplifier node (e.g., SAB) downto a pre-sense write level of 0.9 volts. The column select signal CS.1is then brought low as the sense amplifier begins to latch, followed bythe WR_L or WR_R signal returning low. It is imperative that the columnselect signal is brought low before the WR_L or WR_R signal is broughtlow. This ensures that transistor 404 remains conductive as the columnselect signal (e.g., CS.1) is brought low and is able to discharge node407 fully to VSS, thereby turning off transistors 405 and 406. It isalso imperative that the WR_L or WR_R signal returns to ground beforethe byte write enable signal BW.X is updated with new data, particularlyif the byte write enable signal BW.X is disabled (brought low). If theBW.X signal were brought low before the WR_L or WR_R signal is broughtlow, a high voltage could be trapped on the gate of transistor 404,which could result in an unwanted write during a read cycle. If no writecycle occurs for a long time, the low (inactive) voltage on node 407 ismaintained by both sub-threshold leakage current through transistor 404,and by diode leakage from the drain of transistor 404 (an N+region) toan underlying P-well region biased either at ground or at a voltagebelow ground. Operation of the bit line sense amplifier itself thencompletes the write by driving the high-going bit line up to the actualwrite level of, for this embodiment, about 2.0 volts. Since transistors405 and 406 are off during latch, current into the sense amplifiersduring latch is not communicated to the global input lines.

Although the capacitance of the global input lines is perhaps 20% higherthan other embodiments utilizing a bi-directional local I/O line (as inFIG. 8), the signal on the global input lines is developed by the writequeue before the internal write operation begins and may be accomplishedwithout significant complication. The increase in power consumptionrequired to drive the higher capacitance global input lines isnegligible since the lines are only driven by +/−100 mV. Moreover, thereis only one transistor in the write path from the global input lines tothe sense amplifier nodes. Transistors 405, 406 may therefore beimplemented as smaller devices and still achieve high performance. As isdescribed in detail below, operation at 200 MHz is achievable using thehole circuit embodiments shown in FIG. 10 and FIG. 11.

FIG. 12 is a schematic diagram of an address pre-decoding circuit usefulfor both row and column addresses. Global address signals GAx, GAy, andGAz and their respective complement address signals GAxB (not shown),GAyB, and GAzB (each pulsing high, when active, for approximately 1 ns)are connected in a tree configuration to discharge a selected one-of-Ndecode nodes. For example, global address signals GAx, GAy, and GAz areconnected to the gates of respective transistors 410, 411, and 412 todischarge decode node 414. One of the global address signals (in thisembodiment, GAz) is connected to transistor 413 to precharge the decodenode 414 high when the global address signals are inactive (low). Aself-resetting buffer 415 then inverts and buffers the signal on thedecode node 414 to generate an active high pulsed pre-decoded globaladdress signal GAxyz. Eight such pre-decoded global address signalsGAxyz are generated by pre-decoding three global address signals, withone of the eight pre-decoded signals pulsing high when active. Thevarious row pre-decode signals are advantageously generated (by buffer415) to have a pulse width of slightly more than 3 ns, and the variouscolumn pre-decode signals generated to have a pulse width of slightlymore than 2 ns.

FIG. 13 is a schematic diagram of an address decoder circuit whichreceives global pre-decoded row address lines and generates localpre-decoded row address lines which correspond to and are local to asingle array block within a memory bank. An array decode node 422 withinthe selected array block (which is precharged by PMOS transistor 431when the global pre-decoded address signal GA2.3 is low) is brought toVSS by the series combination of transistors 420, 421 when both theglobal pre-decoded address signal GA15.16 (a pre-decoded signal ofaddress bits 15 and 16) and GA2.3 are high. Since four address bits areencoded in these two pre-decoded signals, a one-of-sixteen decode isaccomplished in selecting the array decode nodes (e.g., node 422). Agroup of eight pre-decoded local address signals LA9.10.11 is generatedby decode circuit 423. One of the eight pre-decoded global addresssignals GA9.10.11 is driven high, and the corresponding pre-decodedlocal address signal LA9.10.11 within the selected array block is drivento follow accordingly. All eight local address signals LA9.10.11 withinother non-selected array blocks remain inactive. A second decodecircuit, labeled 424, generates a group of four pre-decoded localaddress signals LA8.9B which are active low, rather than active high asbefore. Lastly, an additional N-channel decoder tree circuit 425includes four transistors 427 (and four associated PMOS prechargetransistors 430) which each receive a particular one of four pre-decodedglobal address signals GA14.17, with each transistor 427 connected to arespective second tree level which each includes four transistors 426for receiving a particular one of four pre-decoded global addresssignals GA12.13. A precharge transistor 429 and buffer 428 are includedfor each of sixteen pre-decoded local address signals LA12.13.14.17 tocreate active-high signals.

FIG. 14 is a schematic diagram of a row address decoder which receivesVDD-level pre-decoded local address signals, level shifts up to a VPPlevel that is substantially independent of VDD and typically above VDD,and drives a selected word line from VSS to VPP. (It should be notedthat the “body” terminal for all NMOS transistors is tied to VSS, unlessotherwise indicated, and the “body” terminal for all PMOS transistors isnormally tied to VDD, unless otherwise indicated. However, the bodyterminal for any PMOS transistor whose source or drain terminal isconnected to VPP is also connected to VPP. Any inadvertent omission ofthe body terminal for a PMOS transistor whose source or drain isconnected to VPP should not suggest that it is connected to VDD.) Thereare no race conditions within the decoder, even though it accomplishes alevel shifting from VDD-level (i.e., VSS-to-VDD level) pre-decodedaddress signals to VPP-level word lines. A two-level tree includes agroup of eight transistors 445 which each receive a particular one ofeight pre-decoded local address signals LAijk, with each transistor 445connected to a respective second tree level which each includes eighttransistors 444 for receiving a particular one of eight pre-decodedlocal address signals LAlmn. The decode node 441 within the selecteddecoder 440 is precharged to VPP at the end of an active cycle by alevel-shifted version of the local address signal LAlmn, conveyed onnode 453 and connected to the gate of P-channel transistor 442. When thelocal address signal LAlmn goes low at the end of an active cycle, theinverter 459 drives the gate of transistor 458 high. This momentarilyoverpowers P-channel transistor 456 until its gate is pulled high bytransistor 455. Then, node 453 is pulled all the way to VSS to prechargethe decode node 441. When the local address signal LAlmn goes high atthe beginning of an active cycle, the gate of transistor 456 is pulledlow by transistor 457 (briefly overpowering transistor 455) and node 453is pulled at the way to VPP, thereby turning off transistor 442.

It should be noted that transistors 445 are indicated as being highvoltage transistors (as are nearly all others in FIG. 14). From avoltage point of view, transistors 445 do not need to be high voltagetransistors. The maximum voltage of the local address signal LAlmn isVDD, and consequently the maximum voltage on the drains of transistors445 is limited (by transistors 444) to a voltage of VDD less athreshold. Since the maximum voltage of the local address signal LAijkis also VDD, the maximum voltage on the gates of transistors 445 is VDD.Therefore, transistors 445 do not need to be high voltage transistors.However, there may be layout benefits to nonetheless make them highvoltage transistors. High voltage transistors usually requireconsiderable spacing to the nearest regular low-voltage transistor.Since transistors 444 must be high voltage transistors (i.e., its drainvoltage goes to VPP), transistors 445 may be physically closer to thetransistors 444 if both are high voltage transistors, than iftransistors 445 are regular low-voltage transistors. Consequently, thelayout area may be smaller, and the capacitance on the intermediate nodetherebetween may be smaller. Few things in life come for free, however.A high voltage transistor operated at low voltage is lower inperformance than a normal low-voltage transistor operated at lowvoltage, so judicious choices must be made when considering making aparticular transistor a high voltage transistor when the voltageconsiderations do not strictly so require.

All other nodes 441 of other non-selected decoders remain at VPP duringan active cycle. Even though the local address signals LAijk and LAlmnare VSS-to-VDD signals, the N-channel tree configuration is useful todischarge the decode node 441 all the way from VPP down to VSS. Thedecode node 441 within the selected decoder 440 is brought to VSS, whilethe corresponding decode node 441 within all other decoders 440 remainsat VPP. Buffers 446, 447, and 448 are all normally powered by VPP(rather than by VDD) and together invert the active low voltage on theselected decode node 441 and drive the word line 454 smoothly from VSSto VPP.

A redundancy flip-flop 449 is used to disable a particular decoder 440from driving its word line 454. Normally, the flip-flop 449 is resetlow, thus turning on transistor 450 to provide power to buffer 446 andturning off transistor 451 to allow node 452 to switch high (when thedecoder is selected). However, if a redundant word line decoder alsoresponds to the same row address, the signal REDSEL (which is routedthrough all the row decoders within an array block) is driven high, andthe output of flip-flop 449 is driven high to disable buffer 446 and toensure that node 452 remains low. Once redundancy flip-flop 449 is set,the particular word line 454 cannot be driven high and always remainsinactive. The features and advantages of such a redundancy flip-flop aredescribed more fully in co-pending application Ser. No. 09/199,884,filed on Nov. 24, 1998, naming Robert J. Proebsting as inventor andentitled “Disabling a Defective Element in an Integrated Circuit DeviceHaving Redundant Elements,” the disclosure of which is incorporatedherein by reference in its entirety.

The lower power supply connection of buffer 448 is taken to a controlsignal WLGND which during normally operation is driven to VSS by a lowimpedance circuit. However, during power-up (i.e., during an internallycontrolled power-up), the signal WLGND is driven high to VDD for apredetermined amount of time. Consequently, all word lines are drivenhigh by the respective N-channel pull-down transistor within buffer 448(not explicitly shown) which is normally used to “ground” the respectiveword lines. By holding major precharge and array select signals in aprecharge state (in most cases forced to VDD), and by forcing the bitline equilibrate voltage (e.g., VBLEQ shown in FIG. 7) to VSS, allmemory cells are initialized to VSS, as is described in greater detailbelow. Virtually all the transistors in this decoder 43 are indicated asbeing high voltage transistors.

Because the area required for a separate decode node and redundancyflip-flop may preclude such a row decoder 440 matching the word linepitch requirements of a memory array, FIG. 15 shows an embodiment of afinal row decoder 460 for driving four word lines which is conceptuallysimilar to that shown in FIG. 14, but which includes a one-to-fourdecoder in the final word line buffers and two redundancy flip-flopsuseful for replacing pairs of word lines (rather than single wordlines). The particular address signals shown in the figure are, ofcourse, arbitrary but are drawn to match the pre-decoded local addresssignals described in FIG. 13.

In normal operation, each decode node 461 is precharged high to VPP bytransistor 470 whenever the pre-decoded local address signalLA12.13.14.17 is low (inactive). A level-shifting circuit (such as theone which generates node 453 in FIG. 14) may be employed to drive thegate of transistor 470 if the pre-decoded local address signalLA12.13.14.17 is a VDD level signal. Each single decode node 461 servesfour separate word lines WL0, WL1, WL2, and WL3. When one of the fourword lines is to be selected, the local address signal LA12.13.14.17Bconnected to the sources of transistors 468 and 469 is brought low (forthe selected decoder), and the decode node 461 is discharged from VPP toVSS by either transistor 468 or 469 (which has a particular one of theeight local address signals LA9.10.11 coupled to its respective gate).The decode node 461 is then inverted by buffer 462 to generate an activehigh WLEN signal which is driven to VPP. A final one-of-four decode isimplemented by a group of four switched buffers 463, each with aparticular one of the four active-low pre-decoded local address signalsLA8.9B serving as the respective “VSS” connection. A group of fourrespective buffers 464 then drives each respective word line WL0, WL1,WL2, and WL3.

Two redundancy flip-flop circuits 465 and 484 are provided to disablenormal (i.e., non-redundant) word lines in adjacent pairs. In normaloperation, inverters 479 and 480 implement a first flip-flop which isinitialized at power-up so that node 481 is high. Consequently,transistor 478 drives the gate of transistor 466 high to a voltage ofVPP−V_(T) (e.g., drives the gate of transistor 466 to “a threshold belowVPP”). When one of the normally low pre-decoded local address signalsLA9.10.11 is driven high, the gate of transistor 466 bootstraps to passthe full VDD level voltage onto the gate of transistor 468 with littledelay to discharge the decode node 461. However, whenever a firstredundant row decoder responds to a particular row address, and addressbit A9 within the particular row address is a logical “0”, then aredundant row enable signal REDUN0 is driven high. During the firstcycle after powerup which receives this particular row address(including A9=0), the WLEN signal within the selected decoder is drivenhigh. Consequently, transistors 482 and 483 are both conductive sincethe regular decoder is selected (node WLEN is high) and a redundant rowdecoder also selected (signal REDUN0 is high), which discharges node 481(overpowering inverter 480 briefly) and sets the flip-flop (479,480)with node 481 low. As a result, transistor 466 is held off, transistor467 is held on, the gate of transistor 468 held at VSS, and the decoder460 is prevented from ever again responding to this particular rowaddress (with A9=0). The first redundant row decoder is now the onlydecoder ever again responding to that particular row address.

Redundancy flip flop circuit 484 is identical in structure, but receivesa different one of the eight pre-decoded local address signals LA9.10.11(responding to the same A10 and A11 address but the opposite A9 address)and receives a redundant row enable signal REDUN1 from a secondredundant row decoder. The two pre-decoded local address signalsreceived by flip-flops 465 and 484 are chosen to reflect the samepre-decode of bits address 10 and 11 (reflecting a one-of-four decode)but reflecting a complementary address bit 9. This address bit 9 (alongwith address bit 8) is also pre-decoded and forms the basis of theone-of-four decode within buffers 463. The presence of address bit 9within the pre-decoded local address signals LA9.10.11 is a consequenceof layout pitch realities, the use of a single decode node 461 shared byfour word lines, and the desire to replace defective word lines byredundant word lines in groups of two, not in groups of four. (Replacingsingle word lines would be preferred but may be difficult to achieve dueto layout constraints.) Conceptually, one may view pre-decoded localaddress signals LA9.10.11 as a group of four pre-decoded signals (usingA10 and A11) which implement a one-of-four decode if each signal isconnected to a single decoder 460, but wherein each of the four signalsis actually implemented as two signals—one signal reflective of addressbit A9 being a logic 1 (and indicated as LA9.10.11), and the otherreflective of address bit A9 being a logic 0 (which is indicated asLA9#.10.11). Alternatively, if replacing defective word lines in groupsof four is acceptable, the address bit 9 may be removed from thepre-decoded LA9.10.11 signals, and only a single flip-flop circuit 465and a single redundant row enable signal REDUN0 need be used.

FIG. 16 is a schematic diagram of a power-up circuit 500 useful for therow decoder embodiments shown in FIG. 14 and FIG. 15 which duringpowerup turns on all row lines and drives the common bit lineequilibrate node to VSS. During power-up, the control signal PWUBreceived by circuit 500 is low, and consequently the WLGND signal isdriven high by inverter 501 (thus causing all word lines to go high, asdescribed above), transistor 503 is held off, and transistor 504 groundsthe bit line equilibrate line VBLEQ. Since the various equilibratesignals and array select signals are held high during power-up, the bitlines are thus driven to the ground voltage on the VBLEQ node and allmemory cells are written with a “low” (which, of course, may appear aseither a logic “0” or a logic “1” depending on whether the cell isconnected to a true or complement bit line). At the conclusion of thepower-up sequence, the control signal PWUB goes high, large transistor502, turns on to provide a low impedance ground to the final word linebuffer within each row decoder, transistor 504 is held off, andtransistor 503 is turned on and couples an internally generated voltagereceived on VBLEQG (having a typical value of around 1.0 volts) to thebit line equilibrate node VBLEQ.

FIG. 17A is a schematic diagram of a circuit which generates the upperand lower pulsed equilibrate signals and the upper and lower arrayselect signals for a row of sense amplifiers within a given hole betweenarray blocks. Two symmetrical circuits 510 and 511 are shown torespectively generate the upper and lower signals. The operation of bothmay be understood by a description of the upper circuit 510. Betweenactive cycles, the global precharge signal GPRE received by inverter 512is low, and consequently node 514 is high, and the upper pulsedequilibrate signal BLEQU is low. Node 521 was precharged high by anearlier low-going pulse on node 520 (which is now high). Inverter 524drives its output node 518 low, while transistor 519 (whose gate is heldlow by level-shifting inverter 528) is also conductive. Together thesetwo fairly small transistors 519 and 535 weakly maintain the upper arrayselect signal ASU at VDD. Inverter 527 holds transistor 517 off, andlevel-shifting “AND” gate 515 (described in greater detail below) holdstransistor 516 off. As previously noted, the “body” or “swell”connection for all PMOS transistors is tied to VDD, unless otherwiseindicated. As specifically indicated, the body terminal of transistor516 is connected to VPP, rather than its own source potential, VDD, sothat the drain node ASU may be boosted above VDD without forward biasingthe drain-to-body junction.

During an active cycle, global pre-decoded address signals GA2.3D andGA15.16D are received by transistors 522 and 523, respectively, whichrepresent the particular ones of the pre-decoded global address signalsGA2.3 and GA15.16 used to select the array block below the row of senseamplifiers within the given hole. If the array block below the row ofsense amplifiers is to be selected, then transistors 522 and 523 turn onto discharge node 521, inverter 524 drives node 518 high, and inverter518 quickly turns off the upper array select signal ASU. The fallingnode 521 also causes moderately-sized transistor 530 in the lowercircuit 511 to turn on, thus driving the lower array select signal ASDtoward VPP moderately quickly. Moreover, the gate of transistor 531 isdriven high to “arm” inverter 532 to respond to a high-going pulse onthe global precharge signal GPRE.

Near the end of an active cycle as the selected word line is returningto ground, the rising edge of the global precharge signal GPRE arrives,inverter 532 responds, node 520 is driven low, and the lower pulsedequilibrate signal BLEQD is driven high. Transistor 526 is also turnedon, which then quickly turns off the N-channel pull-down transistorwithin inverter 518. At about the same time, level-shifting “AND” gate515 turns on transistor 516 to charge the upper array select signal ASUback up to VDD in preparation for the next cycle. Node 520 going lowalso precharges node 521 high, which then enables the weak currentsource (transistor 519 and the P-channel transistor within inverter 518)which maintains the upper array select signal ASU at VDD. In the lowercircuit 511, node 520 going low and node 521 going high also causestransistor 530 to turn off and transistor 533 to turn on, which drivesthe lower array select signal ASD down from VPP back to VDD. Since node529 was never discharged during this exemplary cycle, transistor 513 isoff and inverter 512 does not respond to the rising edge of the globalprecharge signal GPRE.

When the global precharge signal GPRE goes back low (i.e., at the end ofthe pulse on GPRE), the output of inverter 532 switches high (even morequickly because transistor 531 is already off), node 520 is driven high,and the lower pulsed equilibrate signal BLEQD is terminated (broughtlow). Because node 520 switches back high, the output of level-shifting“AND” gate 515 is driven to VPP and turns off transistor 516. The upperarray select signal ASU had been brought quickly from VSS to VDD by therather large transistor 516 which is now off, and then is weakly held atVDD by transistors 519 and 535 so that it may be discharged quickly atthe start of the next cycle without having to first discharge the gatenode of a large P-channel device, nor without wasting much chargethrough a large inverter during the transition of such a large inverter.NAND gate 525 is used to ensure that the upper array select signal ASUis held high, even if only weakly so, during power-up when the signalPWUB is low. Analogous transistors in all such circuits ensure that allsuch array select signals are also held high during power-up.

FIG. 17B is a schematic diagram of an embodiment of the level-shiftinginverter circuit 528 useful for the circuitry shown in FIG. 17A. Whenthe input signal IN goes low, node 902 is pulled low through transistor901. This momentarily over-powers P-channel transistor 903 until itsgate (output node 906) is pulled high by transistor 904. When the inputsignal IN goes high, the output node 906 is pulled low by transistor 905(briefly over-powering transistor 904). Transistor 901 shuts off andallows cross-coupled P-channel transistor 903 to pull node 902 all theway up to VPP, thereby turning off transistor 904. After brieflyover-powering the respective P-channel transistor upon a transition ofits input signal IN, there is no significant standby current in eitherlogic state.

FIG. 17C is a schematic diagram of an embodiment of the level-shiftingAND-gate circuit 515 useful for the circuitry shown in FIG. 17A. Twocircuit blocks similar to the level-shifting inverter 528 receivingrespective input signals A and B are combined, with a NAND configurationof transistors 915 and 916 (in place of the single transistor 905)pulling the analogous output node 917 down, which is followed by aninverter to generate an overall AND function at the output node 918. Theinternal operation of the level-shifting AND gate 515 is analogous tothat described for the level shifting inverter 528.

FIG. 17D is a schematic diagram of another embodiment of thelevel-shifting AND-gate circuit 515 useful for the circuitry shown inFIG. 17A. Its operation may be easily appreciated by one skilled in theart, given the teaching described above for the level-shifting inverter528 shown in FIG. 17B and the level-shifting AND gate 515 shown in FIG.17C.

FIG. 18 is a block diagram of a preferred embodiment of a VPP generatoruseful for the various circuits described herein, which produces asubstantially fixed voltage, usually above VDD for most process corners,which is referenced to VSS. Before describing the actual circuit, somebackground information about the design requirements of such a VPPcircuit are warranted. In the embodiments described herein, the VPPvoltage is used to drive the selected word line from VSS to VPP, and toraise the two selected array select signals (one just above and one justbelow the selected array block) from VDD to VPP. Since the array selectsignal lines are substantially higher in capacitance than a selectedword line, the charge needed to boost the selected array signal lines isillustrative. At high VDD of, for example, 2.9 volts (assuming a nominalVDD value of 2.5 volts and a nominal VPP value of 4.0 volts) the arrayselect signal line must be boosted from 2.9 volts to 4.0 volts (for atotal of 1.1 volts), whereas at a low VDD of, for example, 2.1 volts,the array select signal line must be boosted all the way from 2.1 voltsto 4.0 volts (for a total of 1.9 volts)—almost twice as much!.

For most two-stage charge pump style circuits having no threshold loss(which circuits are well known in the art), the charge transferred perpump cycle is approximately equal to C·(2·VDD−VPP), where C is thecapacitance of the pump capacitor. If VPP=4.0 volts and VDD=2.9 volts,the charge per cycle is equal to C·(5.8−4.0)=1.8·C. If, instead, VDD=2.1volts, then the charge per cycle is equal to C·(4.2−4.0)=0.2·C. Thus, athigh VDD, the charge provided by a single capacitor pump circuit is 9times that at low VDD. Yet at high VDD, the array select lines need tobe boosted less than at low VDD. Only about 58% as much charge would beneeded at high VDD as at low VDD, but the pump circuit provides ninetimes as much charge at high VDD as at low VDD. A properly sizedcapacitor for low VDD would be 15× larger than a properly sizedcapacitor at high VDD. A capacitor sized to provide the correct chargeat low VDD would generate far too much increase in VPP voltage per pumpcycle at high VDD. Conversely, a capacitor sized to provide the correctcharge at high VDD would only provide about 6% of the charge required atlow VDD.

To prevent these and other problems, the VPP generator 540 shown in FIG.18 includes a plurality of pump circuits 544, each connected to theoutput VPP, and each controlled by a common control circuit 543. Eachsuch pump circuit 544 is enabled to pump according to the amount ofcharge which is needed at a particular time, based on the measured levelof both VDD and VPP. A first regulator 541 compares various fractions ofthe VDD voltage to an internally generated bandgap voltage VBG, while asecond regulator 542 compares various lower fractions of the VPP voltageto the bandgap voltage VBG. Such regulator circuits may be implemented,for example, as a plurality of sensing circuits, each comparing VDD (orVPP) or a resistively-divided fraction thereof to a respective pluralityof reference voltages (each derived from a bandgap voltage or a linearmultiple or fraction thereof). Additional sensing circuits reduces thegranularity of the measurement. Conversely, such regulator, circuits mayalso be implemented, for example, as a plurality of sensing circuits,each comparing a bandgap reference voltage (or a linear multiple orfraction thereof) to a respective plurality of resistively-dividedfractions of VDD and/or VPP.

Both regulators provide outputs to the controller 543. If VDD is low,then more of the pump circuits 544 are enabled for a given cycle. As VDDincreases, fewer such pump circuits 544 are enabled. Similarly, if VPPis particularly low (such as during power-up), then all the pumpcircuits are enabled, while if VPP is already high enough, then none ofthe pump circuits 544 are enabled. In a preferred embodiment, none ofthe pump circuits 544 are enabled if VPP exceeds 4.0 volts, while all ofthe pump circuits 544 are enabled if VPP is less than 3.8 volts. Between3.8 and 4.0 volts, the measured values of both VPP and VDD determine howmany pump circuits are enabled.

For a given VPP and VDD voltage there are a fixed number of pumpsenabled. As VDD increases slightly, the charge per cycle increases, eventhough the same number of pump circuits are enabled, because the VDD isincreasing. However, as VDD further increases slightly, one less pump isenabled, so the charge per cycle abruptly decreases. Then as VDD furtherincreases, the charge per cycle again increases because VDD isincreasing. When plotted as a function against VDD, the charge per pumpcycle thus appears as a sawtooth waveform, which decreases abruptly aseach such pump circuit 544 is successively disabled. The pump circuits544 are preferably not uniformly sized, but instead each size determinedindividually so that the charge per pump cycle, when plotted as asawtooth waveform against VDD, varies from min-to-max as little aspossible over the range of VDD.

A significant amount of internal de-coupling (i.e., filtering)capacitance on the VPP node is provided by the various row decoder andarray select circuits which are unselected during a given cycle. Forexample, the last two buffers within each row decoder provide inaggregate a large effective capacitance. Referring briefly again to FIG.14, an unselected row decoder 440 has node 452 held at ground by buffer446. The P-channel transistor within buffer 447 is biased deeply in thelinear region, and virtually all its channel capacitance, as well as itsgate-to-drain and gate-to-source overlap capacitance, is effectivelybiased with VPP-to-VSS potential across the capacitors. Moreover, theN-channel transistor within buffer 448 has VPP on its gate and holds theunselected word line at VSS, and is therefore also biased deeply in thelinear region. Virtually all of its channel capacitance, as well as itsgate-to-drain and gate-to-source overlap capacitance, serve as filtercapacitors for VPP. In addition, the gate-to-drain overlap capacitanceof both the N-channel transistor within buffer 447 (whose gate is heldto VSS and whose drain is held to VPP) and the P-channel transistorwithin buffer 448 (whose gate is held to VPP and whose drain is held toVSS) provide additional bypass capacitance. Taken together, suchcapacitances provide a significant reservoir of charge on the VPP nodewithout requiring separate devices or structures. For one memory bank(which contains a total of 4096 word lines), the capacitance of the VPPnode to VSS may be about 400 pF. The capacitance of one word line may beabout 1.5 pF and the total capacitance of the two selected array selectlines driven to VPP may be about 7.5 pF. The charge removed from VPP toboost the two selected array select lines from VDD to VPP (nominally 4.0volts) and to boost the selected word line from VSS to VPP may be about20 pC (“pico-coulombs”) at VDD =2.1 volts. Thus, if no charge were addedby the charge pump circuit, one active memory cycle will reduce the VPPvoltage by about 50 mV. The pump circuit is responsible for supplyingabout the right amount of charge to first establish, and then tomaintain, the VPP level at the desired voltage of 4.0 volts.

Nonetheless, VPP may still “wander” in voltage slightly depending on thefrequency and interval between various cycles, in spite of theregulators 541 and 542 within VPP circuit 540. Consequently, a testcontrol signal TEST is received by the control circuit 543 to decreasethe regulated value of VPP by a small amount (e.g., 200 mV) when incertain test modes to ensure reliable operation of the memory devicewhen VPP is actually lower than the minimum expected VPP voltage. Byusing such test modes, adequate operating margins for normal operationmay be more easily assured.

FIG. 19 is a flow chart diagram of a preferred embodiment of thepower-up sequence for initializing all memory cells to a known datastate. At step 550, several actions occur in parallel. The memory cellplate within the memory array is driven toward the desired bit lineequilibration voltage (preferably around 1.0 volts) to established itsvoltage at the eventual bit line equilibration voltage. In doing so thecell plate is driven weakly (being careful to limit the current flowwhich charges the cell plate to an amount less than the output currentof the substrate bias charge pump) to prevent the increasing voltage ofthe memory cell plate and the resultant increasing voltage of the memorycell storage nodes from coupling the substrate positive and causingmassive latchup. The row decoders are overridden to allow every wordline to simultaneously go high. Moreover, the bit line equilibrationvoltage VBLEQ is forced to VSS, and the bit line equilibrate signals andarray select signals are ensured to be high (typically at VDD). Theseactions need not be initiated in any particular order, but all fourpreferably occur simultaneously for a predetermined time. Consequently,since the bit line equilibrate signals are held high and the bit lineequilibration voltage VBLEQ is held low, both true and complementinternal sense amplifier nodes are then coupled to a common node at VSSby precharge (i.e., equilibrate) signals.

Since the array select signals are also held high, each bit line (bothtrue and complement) is thus driven to VSS and all memory cells likewiseare written with a “low” (which, of course, may appear as either a logic“0” or a logic “1” depending on whether the cell is connected to a trueor complement bit line), even if the word lines are only slightly higherthan a threshold voltage above VSS.

At step 551 the override to the row decoders is turned off, and the wordlines are allowed to return to VSS (while ensuring that the array selectsignals and bit line equilibrate signals remain high). This “saves” thewritten low voltage in each memory cell. Then, at step 552, the bit lineequilibrate voltage is driven to its normal level which, for a preferredembodiment, entails driving the VBLEQ signal to its desired level ofabout 1.0 volts (while again ensuring that the array select signals andbit line equilibrate signals remain high). Then, when more normal cyclesbegin, the very first operation in the memory array occurs with memoryarray nodes (bit lines, cell plate) properly established, and all memorycells initialized at one of the two valid states (in this case, a validlow level). The first cycles do not have to try to sense memory cellshaving an initialized voltage near the bit line equilibration voltage,as would likely occur without such a power-up sequence due to couplingfrom the memory cell plate to the memory cells themselves as the memorycell plate reaches its normal level at the bit line equilibrationvoltage. If the voltage of many memory cells were permitted to behalfway between a low and a high level, sense amplifiers attempting tosense and restore such memory cells could be in a meta-stable stateduring sensing for some considerable amount of time. This would decreasethe high level written into the memory cell to a voltage lower thannormal, possibly causing a malfunction.

The first “normal” cycles may actually be internally controlled and partof the power-up sequence. At optional step 553, a series of internal“dummy” cycles (or conditioning cycles) may be performed to morefaithfully initialize any internal nodes not already so initialized, toestablish a bit line equilibrate voltage identical to that resultingfrom normal operation, and to cycle through the row decoders to providean opportunity for any programmed redundant word lines (which may havebeen programmed during manufacture to respond to a particular address)to disable the regular row decoder associated with the defective addressbefore the defective element has a chance to interfere with normaloperation after power-up.

FIG. 20 is a block diagram of an exemplary embodiment of an 18 MBitmemory array 560 with four separate memory banks 561, 562, 563, and 564,having a first dual control block 571 between the first memory bank 561and the second memory bank 562, and having a second dual control block572 between the third memory bank 563 and the fourth memory bank 564.Each memory bank 561, 562, 563, 564 includes thirty-two array blocks,each including 128 horizontally-arranged rows (i.e., word lines) and1152 (1024×9/8) vertically-arranged column's (not including redundantrows and columns). Each column is implemented as a complementary foldedbit line pair, and sense amplifiers and other supporting circuits withinthe holes between array blocks are as described above. Four independentrow decoders, each with its own local address bus, are providedrespectively for the four banks, and are physically arranged in twopairs. The first pair of row decoders are included in the dual controlblock 571 located within a first spline 565 between the left pair ofmemory banks (561, 562), and the second pair of row decoders areincluded in the dual control block 572 located within a second spline566 between the right pair of memory banks (563, 564).

A first group of redundant columns 573 is located adjacent to spline 565at the end of each array block within memory bank 561, and a secondgroup of redundant columns 574 is located adjacent to spline 565 at theend of each array block within memory bank 562. Similarly, a third groupof redundant columns 575 is located adjacent to spline 566 at the end ofeach array block within memory bank 563, and a fourth group of redundantcolumns 576 is located adjacent to spline 566 at the end of each arrayblock within memory bank 564. Each group of redundant columns ispreferably implemented by including an additional block 255 (i.e., the33^(rd) block 255) of 32 complementary bit line pairs, which includestwo sets of global I/O lines, such as is shown in FIG. 4 and FIG. 5.However, unlike the normal (non-redundant) global I/O lines, both of thetwo redundant global I/O lines within each memory bank exit both the topand bottom of the memory array, as shown. A first multiplexer 567 islocated at the top of the memory array 560, which directs (e.g., duringa read cycle) 18-bits of data received from one of the memory banks 561,562, 563, or 564 (representing, for example, the upper half of a 36-bitdata word corresponding to the address of the given cycle) to the datainput/output buffers at the top of the memory array 560. A secondmultiplexer 568 is located at the bottom of the memory array 560, whichsimilarly directs 18-bits of data received from one of the memory banks561, 562, 563, or 564 (representing, for example, the lower half of a36-bit data word corresponding to the address of the given cycle) to thedata input/output buffers at the bottom of the memory array 560.

Within each memory bank (e.g., bank 561), write data conveyed from thetop data multiplexer 567 to the memory bank 561 by thirty-six separatepairs of differential global input lines 578, and write data may beconveyed from the bottom data multiplexer 568 to the memory bank 561 bythirty-six separate pairs of differential global input lines 580. Readdata may be conveyed from memory bank 561 to the top data multiplexer567 by thirty-six separate pairs of differential global output lines577, and read data may be conveyed from memory bank 561 to the bottomdata multiplexer 568 by thirty-six separate pairs of differential globaloutput lines 579. During a read cycle of memory bank 561, either 36-bitsof data (representing a “single word” of data) may be driven onto therespective global output lines (18-bits to the top multiplexer 567,18-bits to the bottom multiplexer 568) for conveyance to the appropriateoutput buffers, or 72-bits of data (representing a “double word” ofdata) may be driven onto the respective global output lines (36-bits tothe top multiplexer 567, 36-bits to the bottom multiplexer 568), 36-bitsfor conveyance to the appropriate output buffers during one cycle, andthe remaining 36-bits for conveyance to the appropriate output buffersduring the next cycle (as is described in greater detail below). Ifthere were no column redundancy the multiplexers 567, 568 would each bean 18×8:1 (i.e., 144:18). But each bank has two redundant global I/Olines (i.e., four physical wires per each global I/O line), each ofwhich can be mapped to serve any of the eighteen data I/O buffers at thetop or any of the eighteen data I/O buffers at the bottom of the memoryarray. Therefore, each memory bank can provide data to a singlehorizontal global data bus line from any of four sources: from acorresponding bit within the upper word (of a 72-bit double word), froma corresponding bit within the lower word (of the same 72-bit doubleword), from a first redundant column, or from a second redundant column.

Input pads for the addresses and various control signals (not shown) aregenerally located in either input pad block 569 to the left of memoryarray 560, or input pad block 570 to the right of memory array 560. Mostdata I/O pads (not shown in FIG. 20) are located at the top or bottom ofthe memory array 560, with some located within input pad blocks 569, 570if sufficient area is unavailable at the top and bottom of the device.

FIG. 21 is a block diagram of memory bank 561, illustrating a preferredarrangement of global output lines (and analogous global input lines,not shown). In an exemplary cycle a selected word line falls withinarray block ARRAY.2. To read the memory cells corresponding to a wordline within the array block ARRAY.2, the bit line sense amplifiers inhole HOLE.1.2 located above the selected array block (labeled as 590)and hole HOLE.2.3 located below the selected array block (labeled as591) must both be activated, which should be readily apparent from theabove descriptions regarding the earlier figures. A full 36-bit word maybe driven onto thirty-six of the seventy-two global output lines byenabling only one of these two holes for driving its global output lines(which saves considerable power). In such a case, eighteen of the bitsare conveyed onto global output lines which exit the memory bank 561 atthe top, while the remaining eighteen bits are conveyed onto otherglobal output lines which exit the memory bank 561 at the bottom The“dots” in FIG. 21 indicate a connection between a local I/O read/writeblock (each serving, for example, a group of sixteen bit line senseamplifiers, as illustrated in FIG. 10) and the global output linepassing overhead. For example, the row of local I/O read/write blockswithin the hole HOLE.1.2 are connected to global output lines 592, 594,596 (and others as indicated). As can be discerned, global output line594 is one of eighteen global output lines which are coupled to a localI/O read/write block within hole HOLE.1.2 and which exit the memory bankat the top, whereas global output line 592 is one of eighteen otherglobal output lines which are coupled respectively to a local I/Oread/write block within hole HOLE.1.2 and which exit the memory bank atthe bottom. As the adjoining table at the top and bottom of FIG. 21makes clear, each of the 36-bits of a single data word (i) is conveyedeither to the top or bottom of the memory bank 561 by enabling only theglobal output circuits within a single hole HOLE.1.2. In particular,bits 0-8 of byte 0 and bits 0-8 of byte 1 are conveyed to the bottom ofthe memory bank, while bits 0-8 of byte 2 and bits 0-8 of byte 3 areconveyed to the top of the memory bank. Conversely, enabling only theglobal output circuits within the single hole HOLE.2.3 conveys adifferent 36-bit data word (i+1) from the memory array. Nonetheless,bits 0-8 of byte 0 and bits 0-8 of byte 1 of this data word (i+1) areconveyed to the bottom of the memory bank, while bits 0-8 of byte 2 andbits 0-8 of byte 3 of this data word (i+1) are conveyed to the top ofthe memory bank.

Conversely, two full 36-bit words may be driven onto seventy-twocorresponding global output lines by enabling the read amplifiers inboth of the two holes above and below the selected array block fordriving its global output lines (e.g., by driving in both holes theselected one-of-eight column select signals CS.X, the selected one ofthe left/right read select signals READ_L or READ_R, and the non-decodedREAD signal, all shown in FIG. 10, which traverse through each hole).

As in the case above, and in all cases, both holes are also enabled forbit line sense amplifier sense/restore. In this case, eighteen bits fromthe first word and eighteen bits from the second word are conveyed ontoglobal output lines which exit the memory bank 561 at the top, while theremaining eighteen bits from the first word and the remaining eighteenbits from the second word are conveyed onto other global output lineswhich exit the memory bank 561 at the bottom. The row of local I/Oread/write blocks within the hole HOLE.2.3 are connected to globaloutput lines 593, 595, 597 (and others as indicated). Referring again tothe adjoining table at the top and bottom of FIG. 21, each of the36-bits of the second data word (i+1) is conveyed either to the top orbottom of the memory bank 561 by enabling the global output circuitswithin hole HOLE.2.3. In particular, as before, bits 0-8 of byte 0 andbits 0-8 of byte 1 are conveyed to the bottom of the memory bank, whilebits 0-8 of byte 2 and bits 0-8 of byte 3 are conveyed to the top of thememory bank.

While the assignment of bit order within a 36-bit or 72-bit parallelaccess of the memory bank 561 is potentially arbitrary, the assignmentsshown ensure that a 36-bit single word may be accessed by enabling thelocal I/O read/write blocks within only one hole to drive its globaloutput lines, and still route 18 of the bits to the top of the memoryarray (where half the data output buffers are located) and route theremaining 18 bits to the bottom of the memory array (where the otherhalf the data output buffers are located). Of the 36 global output linesexiting the memory array at the bottom, the 18 bits of each word couldhave been assigned in two contiguous groups, but rather the assignmentshown, where the corresponding bits off both word (i) and word (i+1) areadjacent, is preferred. As an example, global output lines 592 and 595exit at the bottom as adjacent global output lines, and are assigned asbit 0, byte 0, of respective words (i) and (i+1). All 72 global outputlines are assigned, as they exit the memory array, as adjacent pairsmapped to the same bit and byte of the two words (i) and (i+1). Each ofthe memory banks 561, 562, 563, and 564 are preferably arrangedidentically.

FIG. 22 is a block diagram of the memory array within the embodimentshown in FIG. 20 using the arrangement of global I/O lines as shown inFIG. 21 to reduce the worst case length of an internal horizontal databus. Each memory bank 561, 562, 563, 564 is served by a respectivemultiplexer portion 568A, 568B, 568C, 568D, each of which incorporatesan 18-bit×2:1 multiplexer (excluding the redundant columns). The firsttwo global output lines 592, 595 exiting the bottom of memory bank 561are shown, one of which is conveyed onto global data bus GDB.0 bymultiplexer 568A when memory bank 561 is enabled. Likewise, the firsttwo global output lines 592, 595 exiting the bottom of respective memorybanks 562, 563, and 564 are also shown, one of which is conveyed ontoglobal data bus GDB.0 by respective multiplexers 568B, 568C, or 568Dwhen the respective memory bank 562, 563, or 564 is enabled.

Global data bus GDB.0 is associated with a data input/output bufferDATA.0 which may be located not at the bottom of the device, but rather“around the corner” and up the left side of the device. Consequently,the length of the global data bus GDB.0 extends for some distance up theleft side of the device to reach its buffer DATA.0. If each of theeighteen global data buses extended fully across all four memory banks,global data bus GDB.0 would be longer than with this preferredorganization. As shown, the global data bus GDB.0 need not extend acrossthe bottom of memory bank 564 more than necessary to reach the first twoglobal output lines 592,595, since these two global output linescorrespond to bit 0, byte 0 for both data words (i) and (i+1).

Similarly, global data bus GDB.17 is associated with a data input/outputbuffer DATA.17 which is located “around the corner” and up the rightside of the device. Consequently, the length of the global data busGDB.17 extends for some distance up the right side of the device toreach its buffer DATA.17. As shown, the global data bus GDB.17 need notextend across the bottom of memory bank 561 more than necessary to reachthe right-most two global output lines 610,611, since these two globaloutput lines correspond to bit 8, byte 1 for both data words (ii) and(i+1).

Using this bit assignment of global I/Os, each global data bus onlyneeds to extend for a length of slightly more than three of the memorybanks (rather than all four), plus any additional distance necessary toreach its associated buffer. Moreover, the length of the global databuses is reduced even if all the buffers DATA.x are located below thememory banks (e.g., representative global data bus GDB.8 and associatedbuffer DATA.8) and none are located around the corner and up the side ofthe memory array.

FIG. 22B is a schematic block diagram of an embodiment of the data pathbetween the global I/O lines and the external data output pin whichincorporates differential global data bus lines traversing horizontallyacross the top and bottom of the memory device. In the figure, one suchhorizontal differential global data bus GDB.X, GDBB.X is shown, which isrepresentative of each of the eighteen global data buses traversinghorizontally across the bottom of the memory device. (An analogousstructure is found at the top of the memory array, as well, to serveeach of the eighteen global data buses traversing horizontally acrossthe top of the memory device.)

Two non-redundant global I/O interfaces 1060, 1061, and two redundantglobal I/O interfaces 1062, 1063 couple respective global I/O lineswithin memory bank 561 to/from the global data bus GDB.X, GDBB.X. Forthe particular one of the eighteen data bits the global data bus GDB.X,GDBB.X corresponds to, the non-redundant global I/O interface 1060couples the non-redundant global I/O line corresponding to a bit withinthe lower word (i.e., “word 0”) of a 72-bit double word (e.g., globalI/O line 592 shown in FIG. 21), and the non-redundant global I/Ointerface 1061 couples the non-redundant global I/O line correspondingto a bit within the upper word (i.e., “word 1”) of the 72-bit doubleword (e.g., global I/O line 595 shown in FIG. 21). Similarly, theredundant global I/O interface 1062 couples the redundant global I/Oline corresponding to a bit within the lower word and the redundantglobal I/O interface block 1061 couples to the redundant global I/O linecorresponding to a bit within the upper word. (both of which arecollective labeled 573, shown in FIG. 20).

Four such global I/O interfaces are included in each of the other threememory banks. In the adjacent memory bank, two non-redundant global I/Ointerfaces 1066, 1067, and two redundant global I/O interfaces 1064,1065 couple respective global I/O lines within memory bank 562 to/fromthe global data bus GDB.X, GDBB.X. For the particular one of theeighteen data bits the global data bus GDB.X, GDBB.X corresponds to, thenon-redundant global I/O interface 1066 couples the non-redundant globalI/O line corresponding to a bit within the lower word (e.g., analogousto global I/O line 592 shown in FIG. 21), and the non-redundant globalI/O interface 1067 couples the non-redundant global I/O linecorresponding to a bit within the upper word (e.g., analogous to globalI/O line 595 shown in FIG. 21). Similarly, the redundant global I/Ointerface 1064 couples the redundant global I/O line corresponding to abit within the lower word, and the redundant global I/O interface block1065 couples the redundant global I/O line corresponding to a bit withinthe upper word (both of which are collective labeled 574, shown in FIG.20).

The global data bus GDB.X, GDBB.X extends to the other two memory banks,563, 564, as indicated. Although not shown in FIG. 22B, within each ofthe memory banks 563, 564, two non-redundant global I/O interfaces, andtwo redundant global I/O interfaces couple respective global I/O lineswithin each respective memory bank to/from the global data bus GDB.X,GDBB.X. A pair of P-channel “load resistor” transistors 1068, 1069biases each global data bus near VDD when active, and pulls each of theglobal data bus lines to VDD when the global data bus is inactive. Adata I/O interface 1080 (analogous to data input/output buffer DATA.X inFIG. 6) couples the global data bus GDB.X, GDBB.X to/from the particularexternal data pin 1090 (analogous to data pin PAD.X in FIG. 6) for theparticular bit represented.

The operation of each of these global I/O interfaces may be appreciatedby describing the operation of the non-redundant global I/O interface1066, whose major internal components are shown, and its interactionwith the data I/O interface 1080. When signal first develops on theglobal output lines GOUT, GOUTB in a read cycle, it is immediatelybuffered by amplifier 1071 (which is enabled by an “enable read” signal,ENR) to develop a signal on the global data bus lines. This allows thesignal, as it develops, to propagate onto the global data bus lineswithout requiring a carefully timed latch signal. Then, after additionalsignal has developed on the global output lines GOUT, GOUTB, and nearthe peak of its signal (and importantly, before the signal starts todisappear after the read amplifiers (e.g., 372 in FIG. 10) are turnedoff) a latching buffer 1072, also connected to the global output linesGOUT, GOUTB is strobed by a “latch read” signal, LR, to latch the data.The latching buffer 1072 saves this data in the case of a burst moderead cycle. When the burst mode read cycle later occurs, rather thanexecuting a memory array cycle to retrieve the other half of the doubleword (in this example, the lower word), the data is already present inthe latching buffer 1072 from the previous 72-bit “load read” cycle, anddriver 1073 is enabled by an “enable latched read” signal ENLR to drivethe latched data onto the global data bus, and to the data I/O interface1080. In a burst mode read cycle, the “enable latched read” signal ENLRis enabled at the same relative time in the cycle as the “enable read”signal ENR is enabled in a load read cycle.

The differential signal developed on the global data bus GDB.X, GDBB.X,whether driven by the amplifier 1071 or by driver 1073, is approximately100 mV, biased near VDD, and lasts for only 1-2 ns. It is then amplifiedby an amplifier 1081 within the data I/O interface 1080, which developsa 300-400 mV signal on its complementary output nodes 1091 which is alsobiased near VDD. The description thus far is appropriate for both theflow-through mode of operation and the pipelined mode of operation.

In the flow-through mode of operation, this signal is steered throughmultiplexer 1082 (selecting the “B” input) to a latching amplifier 1083.When strobed by a 1-2 ns “latch output” signal LQ, the latchingamplifier 1083 amplifies the read data signal received from themultiplexer 1082 and briefly drives one of its complementary outputnodes active (in accordance with the polarity of the data signal, ofcourse). This brief pulse on one of the two complementary outputs oflatching amplifier 1083 drives a respective input of a latching outputbuffer 1084. This latching output buffer 1084 latches the data andimmediately drives the external data pin 1090 accordingly. The latchingoutput buffer 1084 remains latched, and continues to drive the externaldata pin 1090, until the data is updated when a subsequent LQ pulsestrobes the latching amplifier 1083 with the next data signal (unlessthe latching buffer 1084 is disabled by a control signal which turns offboth pull-up and pull-down drivers within the latching output buffer1084, which occurs when a “stop” or a “load write” command follows aread in progress). In the flow-through mode of operation, the latchingbuffer 1085 and registers 1086, 1087 remain inactive.

In the pipelined mode of operation, the differential signal developed onthe global data bus GDB.X, GDBB.X, whether driven by the amplifier 1071or by driver 1073, is again approximately 100 mV, biased near VDD, andlasts for only 1-2 ns. It is amplified by an amplifier 1081 within thedata I/O interface 1080, which eventually develops a 300-400 mV signal(also biased near VDD) on its complementary output nodes 1091 (see, forexample, nodes DOUT, DOUTB in FIG. 28). Early in the active read cycle,the multiplexer 1082 is configured (selecting the “B” input) to couplethe complementary nodes 1091 to the complementary output nodes 1094 ofthe multiplexer 1082, so that the data signal is available as soon aspossible at the latching amplifier 1083, even before reaching its peak.However, near the peak of this data signal on complementary nodes 1091(and importantly, before it, too, starts to disappear) a latching buffer1085 is strobed by a “latch global data bus” signal, LGDB, to latch thisdata onto its complementary output nodes 1092. The latched signal oncomplementary nodes 1092 is a short pulse on either the true orcomplement node. This pulse is steered into one of two registers 1086,1087, whose complementary outputs are rail-to-rail signals. (The highoutput is essentially at VDD, and the low output is essentially atground.)

Assume for a moment that the data signal is loaded into register 1086.Consequently, the data signal from the amplifier 1081 is available firstat the “B” input of the multiplexer 1082, and last for 1-2 ns. Butbefore it goes away, the register 1086 latches the same data signal,which is then available at the “C” input of the multiplexer 1082 (ascomplementary rail-to-rail signals, as described above). The multiplexer1082 is switched from its “B” input to its “C” input after the “C” inputdata signal becomes available but before the “B” input data signal goesaway, so that the data signal is continuously available at inputs of thelatching amplifier 1083. By using this arrangement, the data signal ismade available to the latching amplifier 1083 as soon as possiblewithout having to wait for the signal to develop sufficiently to strobelatching buffer 1085. Yet the same data signal remains available at thelatching amplifier 1083 even after the signal on the global data busGDB.X, GDBB.X and the signal on the complementary output nodes 1091 hasgone away!

The latching amplifier 1083 is strobed by a 1-2 ns “latch output” signalLQ to amplify the read data signal received from the multiplexer 1082and briefly drive one of its complementary output nodes active (inaccordance with the polarity of the data signal, of course). As before,this brief pulse on one of the two complementary outputs of latchingamplifier 1083 drives a respective input of a latching buffer 1084,latches the data and immediately drives the external data pin 1090accordingly. Assuming the data outputs are driven at a time ⅓ of the wayinto the next external clock period, the LQ signal may occur very earlyin the next active cycle (for very fast external clock cycle times), ormay occur well after the entire active cycle is complete and the memorydevice has automatically returned to an equilibrated state (for veryslow external clock cycle times). The two registers 1086, 1087 areprovided to account for this widely varying arrival time (relative tothe internal active cycle timing) of the LQ signal which strobes thelatching amplifier 1083.

Assume for moment the memory device is operating in the pipelined mode,with an external clock cycle time of 20 ns. The data output pins areconsequently driven at approximately 7 ns into the next cycle. In otherwords, the data signal from the last read cycle, which may be stored inregister 1086, is strobed into the latching amplifier 1083 and (by wayof the latching buffer 1084) driven onto the external data pin 1090 at atime 7 ns after the current cycle is initiated. If the data signal fromthis current cycle arrives at the data I/O interface 1080 before thistime, it is stored (before it goes away) in the second register 1087,because the data in register 1086 (from the last cycle) has not yet beenconveyed to the latching amplifier 1083. At about 7 ns, the data signalfrom register 1086 is strobed by the latching amplifier 1683, afterwhich the multiplexer 1082 is switched from its “C” input to its “D”input so that the data signal conveyed through the multiplexer 1082 andto the latching amplifier 1083 during the following cycle is the readdata signal from the current cycle.

Conversely, assume for moment the memory device is operating, in thepipelined mode at a very fast external cycle time of 5 ns. The dataoutput pins are consequently driven at approximately 1.7 ns. Furtherassume that the internal “access time” of the memory device results inthe data signal reaching the data I/O interface 1080 at about 6 ns afterthe start of its cycle. In other words, the data signal from the lastread cycle reaches the amplifier 1081 only 0.7 ns before it needs to bestrobed into the latching amplifier 1083 and driven (by way of thelatching buffer 1084) onto the external data pin 1090. There isinsufficient time to wait until the data signal is loaded into aregister before presenting it to the latching amplifier 1083.Consequently, the “B” input of the multiplexer 1082 is selected toconvey the data signal directly from the amplifier 1081 to themultiplexer 1082. Then, latter in the active cycle, after the same datasignal is also stored into one of the two registers 1086, 1087, themultiplexer 1082 is switched from its “B” input to either its “C” or “D”input so that the same data signal is still available to the latchingamplifier 1083. In this way, the proper data signal is conveyed by themultiplexer 1082 to the latching amplifier 1083, independently ofwhether the LQ signal which strobes the latching amplifier 1083 arrives“early” or “late” in the cycle.

In the flow-through mode, the LQ signal timing is preferably made moreor less aggressive in the fashion described elsewhere herein regardingthe configurable t₄ timing signal. In the pipelined mode, the LQ signalis instead driven by the last of two events to occur. First, the datamust already be available. That is, the timing signal t₄ from theprevious clock cycle must have already occurred. And second, theappropriate time to output data (e.g., at ⅓ of the way into the presentexternal clock cycle) has already occurred. If the timing signal t₄ fromthe previous cycle has not yet occurred at a time ⅓ of the way into thepresent cycle, the LQ signal must be delayed until the data signal fromthe previous cycle is present (as a result of the arrival of timingsignal t₄).

For clarity, the data input path is also shown in the figure, althoughthe operation details and relative timing is largely described elsewhereherein. Within the data I/O interface 1080, a data input buffer 1088, ifenabled by an ENABLE signal, and upon the rising edge of the internalclock signal CLK, strobes the external data signal presented to theexternal data pin 1090 (comparing it with an internally generated VRRreference voltage), and conveys the data signal to the write queue 1089(the data portion thereof). A read bypass path is provided for, andselected by the “A” input of the multiplexer 1082, to provide data,during a read cycle, which addresses the same word as an earlier writestill pending in the write queue which has not yet been retired bywriting the corresponding memory cells in the array.

Within the global I/O interface 1066, a differential latching buffer1074 receives the small-signal differential data input signal whendriven by the write queue 1089 onto the global data bus GDB.X, GDBB.X,which is then conveyed to global input line pulse driver 1075 to drivethe +/−100 mV signal onto the selected GIN, GINB lines.

Generally, the linear amplifiers or drivers shown in FIG. 22B may beimplemented similarly to the GOUT amplifier within circuit block 372(shown in FIG. 10), the various latching amplifiers shown in FIG. 22Bmay be implemented similarly to the amplifier 193 (shown in FIG. 9), andthe global input line pulse driver 1075 may be implemented similarly todriver 190 (shown in FIG. 3 and FIG. 9).

FIG. 23 is a block diagram of a memory bank (e.g., memory bank 561) inaccordance with another embodiment of the invention which arranges theglobal I/O lines (i.e., the global input lines and the global outputlines) so that all nine bits within each byte are contiguous to eachother. In the figure, only the global output lines are shown, one drawnline representing a differential pair (like in FIG. 21), and each globalinput line pair (not shown) runs substantially adjacent to thecorresponding global output line pair. In an exemplary cycle a selectedword line falls within array block ARRAY.1. To read the memory cellscorresponding to a word line within the array block ARRAY.1, the bitline sense amplifiers in hole HOLE.0.1 located above the selected arrayblock and hole HOLE.1.2 located below the selected array block must bothbe activated. A full 36-bit word may be driven onto thirty-sixcorresponding global output lines by enabling only one of these twoholes for driving its global output lines. In such a case, eighteen ofthe bits are conveyed onto global output lines which exit the memorybank 561 at the top, while the remaining eighteen bits are conveyed ontoother global output lines which exit the memory bank 561 at the bottom.The “dots” in FIG. 23 indicate a connection between a local I/Oread/write block (each serving, for example, a group of sixteen bit linesense amplifiers) and the global output line passing overhead. Forexample, the row of local I/O read/write blocks within the hole HOLE.0.1are connected to the first nine odd-numbered global output lines 621(corresponding to Byte 3, Word 1) which exit the memory bank at the top,the second nine even-numbered global output lines 622 (corresponding toByte 0, Word 1) which exit the memory bank at the bottom, the third nineodd-numbered global output lines 623 (corresponding to Byte, 2, Word 1)which exit the memory bank at the top, and the fourth nine even-numberedglobal output lines 624 (corresponding to Byte 1, Word 1) which exit thememory bank at the bottom. Consequently, each of the 36-bits of a single36-bit data word is conveyed either to the top or bottom of the memorybank 561 by enabling only the global output circuits within a singlehole HOLE.0.1.

Conversely, two fill 36-bit words may be driven onto seventy-twocorresponding global output lines by enabling both of the holes aboveand below the selected array block for driving its global output lines.(As in the case above, both holes are also enabled for bit line senseamplifier sense/restore.) In this case, eighteen bits from the firstword and eighteen bits from the second word are conveyed onto globaloutput lines which exit the memory bank 561 at the top, while theremaining eighteen bits from the first word and the remaining eighteenbits from the second word are conveyed onto other global output lineswhich exit the memory bank 561 at the bottom. For example, when bothholes HOLE.0.1 and HOLE.1.2 are enabled for read/write (e.g., drivingits global output lines when a read cycle), the first nine odd-numberedglobal output lines 621 (corresponding to Byte 3, Word 1) which exit thememory bank at the top are driven by read circuitry (e.g., a local I/Oread/write block) in HOLE.0.1, the second nine odd-numbered globaloutput lines 628 (corresponding to Byte 3, Word 0) which exit the memorybank at the top are driven by read circuitry in HOLE. 1.2, the thirdnine odd-numbered global output lines 623 (corresponding to Byte 2, Word1) which exit the memory bank at the top are driven by read circuitry inHOLE.0.1, and the fourth nine odd-numbered global output lines 629(corresponding to Byte 2, Word 0) which exit the memory bank at the topare driven by read circuitry in HOLE.1.2.

While the assignment of bit order within a 36-bit or 72-bit parallelaccess of the memory bank 561 is potentially arbitrary, the arrangementshown ensures that a 36-bit word may be accessed by enabling the localI/O read/write blocks within only one hole to drive its global outputlines, and still route 18 of the bits to the top of the memory array(where half the data output buffers are located) and route the remaining18 bits to the bottom of the memory array (where the other half the dataoutput buffers are located). By arranging the nine global output linesfor a single byte as a contiguous group, the byte write enable circuitrymay be conveniently located with each group, particularly if routedvertically from a circuit either at the top or bottom, as appropriate.Each of the memory banks 561, 562, 563, and 564 of FIGS. 20 and 22 mayalternatively be arranged as shown in FIG. 23.

FIG. 24 is a block diagram of a portion of a memory bank as illustratedin FIG. 21 (and equally valid for the memory bank shown in FIG. 23)diagramming which array signals are active or inactive for both a normalcycle addressing a selected word line in an arbitrary array block (only36-bits driven onto global output lines), as well as for a burst readcycle (72-bits driven onto global output lines) or a merged write cycle(up to 72-bits written within the same cycle). Three arbitrary arrayblocks ARRAY.W, ARRAY.X, and ARRAY.Y are shown, along with correspondingholes HOLE.V.W, HOLE.W.X, HOLE.X.Y, and HOLE.Y.Z. For the exemplarycycle described, assume the selected word line falls within array blockARRAY.X, as shown, and the addressed 36-bit word corresponds to globalinputs and outputs connected to read/write circuits within holeHOLE.W.X.

Since the selected word line falls within array block ARRAY.X, bothholes HOLE.W.X and HOLE.X.Y are enabled for bit line sense amplifiersense/restore of array block ARRAY.X. Looking first at the signalswithin hole HOLE.W.X, the upper array select signal ASU (i.e., the“unselected array select” for this exemplary cycle) is brought to VSS toisolate the selected sense amplifiers from the bit lines within arrayblock ARRAY.W. The upper bit line equilibrate signal BLEQU (for the bitlines in array block ARRAY.W) remains at VSS to save power. The lowerarray select signal ASD (i.e., the “selected array select” for thisexemplary cycle) is boosted above VDD to provide a lower impedance pathbetween the selected sense amplifiers and the corresponding bit lineswithin the selected array block ARRAY.X. The lower bit line equilibratesignal BLEQD is already at VSS which allows signal to develop on the bitlines within the selected array block ARRAY.X. At the end of the activecycle, the ASD signal is brought back to VDD and the BLEQD signal ispulsed. To accomplish the sense/restore, the sense amplifier equilibratesignal SAEQ_LEVEL is brought to VSS to allow signal to develop on theinternal sense amplifier nodes, the complementary sense amplifier enablesignals SE, SEB are pulsed at the appropriate time and duration(described in detail elsewhere herewithin) to restore the high and lowlevels on the bit lines, then both the sense amplifier equilibratesignals SAEQ_LEVEL and SAEQ_PULSE, and the BLEQD signal, are brought toVDD. Lastly the SAEQ_PULSE and BLEQD signals are then brought back lowto VSS. In the FIG. 24, each of these signals is indicated as being“ACTIVE.”

Since the hole HOLE.W.X is also enabled for read/write, the variouscolumn circuits and read/write circuits are also enabled to drive itscorresponding global output lines or to receive data signals from itscorresponding global input lines (for those which are enabled for bytewrite). The column select signals CS.X are enabled, and the selectedone-of-eight is driven high. For a read cycle, the selected one ofeither the READ_L or READ_R signal is driven high (and any non-decodedREAD signal). Conversely, for a write cycle the selected one of eitherthe WRITE_L or WRITE_R signal is driven high (and any non-decoded WRITEsignal which may be employed). The four byte write signals BYTEW areeach previously driven to VDD to enable write to its corresponding byte,or driven to VSS to disable write to its corresponding byte. Since mostcycles typically write all four bytes, the four byte write signalsusually remain at VDD. However, just after a write cycle, the four bytewrite signals for the next write cycle (even if many read cycles areperformed before the next write cycle) are driven as appropriate withinthe particular hole which will be selected during the next write cycle,the next address and data to be written already presented to the memorydevice and stored within the write queue.

Looking next at the signals within hole HOLE.X.Y (which is also enabledfor bit line sense amplifier sense/restore of array block ARRAY.X), thelower array select signal ASD (i.e., the “unselected array select” forthis exemplary cycle) is brought to VSS to isolate the selected senseamplifiers from the bit lines within array block ARRAY.Y. The lower bitline equilibrate signal BLEQD remains at VSS to save power. The upperarray select signal ASU (i.e., the “selected array select” for thisexemplary cycle) is boosted above VDD to provide a lower impedance pathbetween the selected sense amplifiers and the corresponding bit lineswithin the selected array block ARRAY.X. The upper bit line equilibratesignal BLEQU is already at VSS which allows signal to develop on the bitlines within the selected array block ARRAY.X. At the end of the activecycle, the ASD signal is brought back to VDD and the BLEQU signal ispulsed. To accomplish the sense/restore, the sense amplifier equilibratesignal SAEQ_LEVEL, the complementary sense amplifier enable signals SE,SEB, and the pulsed sense amplifier equilibrate signal SAEQ_PULSE areeach active, and behave as described above.

In a normal 36-bit internal cycle, the hole HOLE.X.Y (unselected in thisexemplary cycle) is not enabled for read/write. In such a cycle, thevarious column circuits and read/write circuits are not enabled to driveits corresponding global output lines or to receive data signals fromits corresponding global input lines (for those which are enabled forbyte write). In particular, the column select signals CS.X, the READ_Land READ_R signals (and any non-decoded READ signal which may beemployed), the WRITE_L and WRITE_R signals (and any non-decoded WRITEsignal which may be employed) are all inactive and remain at VSS. At theend of a write cycle the four byte write signals BYTEW.X within thehole(s) which will be write enabled (i.e., selected) during the nextinternal write operation are, driven to reflect which of those bytes areto be written. If all four bytes were enabled on all write cycles, thenall four byte write signals in all holes would remain at VDD throughoutall cycles.

Alternatively, in a burst read cycle 72-bits are driven onto respectiveglobal output lines, and in a merged write cycle up to 72-bits arewritten with respective data signals on respective global input lines.In such a cycle requiring a simultaneous 72-bit internal access to thememory array, the various column circuits and read/write circuits withinthe hole HOLE.X.Y are also enabled for read/write. As is, the case forhole HOLE.W.X, the column select signals CS.X are enabled (the selectedone-of-eight driven high), the selected one of either the READ_L orREAD_R signal (as well as any non-decoded read signal) is driven high(for a read cycle), and the selected one of either the WRITE_L orWRITE_R signal (as well as any non-decoded write signal) is driven high(for a write cycle). The four byte write signals BYTEW.X each weredriven at the end of the most recent write cycle (for non-merged writecycles) by the write queue to VDD or VSS to accordingly enable ordisable the corresponding byte for the next (i.e., this) write cycle, asdescribed above. Since the byte write signals for a single 36-bit wordare driven at the end of a given write cycle, and since writing a single36-bit word requires enabling only one hole, the four byte write signalswithin a single hole are driven at the end of a given write cycle. Twosuccessive write cycles are required to drive the eight total byte writesignals within the two selected holes prior to a merged write. Then, atthe end of the merged write cycle the four byte write signals BYTEW.Xwithin the hole which will be write enabled (i.e., selected) during thenext internal write operation are driven to reflect which of those bytesare to be written. If all four bytes are to be enabled during the nextwrite operation and were all enabled to be written during the last writeoperation when this hole was selected, then all four byte write signalsBYTEW.X remain at VDD. All other byte write signals within other holesremain unchanged, as already described.

Up to this point, the two holes adjacent to the selected array blockhave been described. All other holes (represented here by holes HOLE.V.Wand HOLE.Y.Z) are non-selected and thus inactive, meaning neitherenabled for bit line sense/restore nor for read/write. Consequently,both the upper and lower array select signals ASU and ASD remain at VDDto continue coupling the sense amplifiers to the adjoining bit lines.The upper and lower bit line equilibrate signals BLEQU and BLEQD alsoremain at VSS to save power. The sense amplifier equilibrate signalSAEQ_LEVEL remains at VDD to continue equilibrating the internal senseamplifier nodes and the bit lines coupled thereto, the complementarysense amplifier enable signals SE, SEB remain at their respectiveinactive levels of VSS and VDD, and the pulsed sense amplifierequilibrate signals SAEQ_PULSE remains at VSS (without pulsing). In theFIG. 24, each of these signals so described is indicated as being“INACTIVE.” The column select signals CS.X, the READ_L and READ_Rsignals (and any non-decoded READ signal which may be employed), theWRITE_L and WRITE_R signals (and any non-decoded WRITE signal which maybe employed) are also all inactive and remain at VSS. The four bytewrite signals BYTEW within this hole behave as described above, which isindicated as “QUIESCENT.” That is, the four byte write signals withinthe hole to be selected for the next write operation are driven toenable (VDD) or disable (VSS) these bytes as required. The four byte,write signals in the presently selected hole remain unchanged at the endof the present write operation (unless the same hole is also selectedfor the next write operation with different bytes enabled). The fourbyte write signals in all other holes remain unchanged.

FIG. 25 through FIG. 31 are waveform diagrams illustrating the majorarray and sense amplifier signals described above, based upon circuitsimulations. For ease of description, general terminology introducedthus far for a selected array block affords adequate reference tospecific signals and description without requiring specific reference toa specific one of the array blocks. It should be readily understood thatthe descriptions which follow are appropriate for any of the arrayblocks when selected. Each of the figures depicts a read or write cycleinitiated by a positive transition of the external clock (not shown) att=19 ns. Earlier cycles establish realistic voltages on the variousnodes at the start of this cycle shown.

Referring now to FIG. 25, the waveforms are shown for a read cycle,sensing and restoring a high from the selected memory cell whileoperating at a VDD of 2.9 volts. Before the start of the cycle shown,the bit lines (not shown) and internal sense amplifier nodes SA, SAB areequilibrated together at a voltage of about 1.2 volts (i.e., the “bitline equilibration voltage”). The upper and lower array select signalsfor the bit line sense amplifiers within a given hole are both inactiveat VDD (one of which will be decoded during the cycle shown as the“selected” array select signal and the other decoded as the “unselected”array select signal). The memory cell to be accessed is shown with astored high level of approximately 2.2 volts.

At t=19 ns, a rising edge of the external clock (not shown) initiatesthe active cycle shown. At about t=21.5 ns, the unselected array selectsignal is quickly brought to VSS to decouple the selected senseamplifier from the bit lines in the adjoining non-selected array block,while the selected array select signal is boosted moderately quickly tothe VPP voltage (which is internally generated and regulated to a valueof about 4.0 volts above VSS) to better couple the selected senseamplifier to the bit lines in the adjoining selected array block (bylowering the impedance of the array select transistors coupling the bitlines to the internal nodes of the bit line sense amplifiers. The senseamplifier equilibrate signal SAEQ_LEVEL (not shown) is also quicklybrought to VSS at substantially the same time (i.e., with the “falling”unselected array select signal) to turn off the lateral equilibrationtransistor in the sense amplifier. Shortly thereafter, the selected wordline is very quickly driven from VSS to VPP and the decoded columnselect signal (e.g., CS.X in FIG. 10) is very quickly driven from VSS toVDD. It is important for the sense amplifier equilibrate signalSAEQ_LEVEL and the unselected array select signal to go low (actually,to go below the bit line equilibration voltage plus a threshold voltage)before the word line is high enough (a threshold voltage above VSS) toimpart any signal from the selected memory cell onto the bit lines andthe internal sense amplifier nodes.

Shortly after the high-going selected word line voltage exceeds the bitline equilibrate voltage plus a threshold (i.e., at about t=22 ns), thememory cell access transistor begins to turn on (the selected memorycell storing a high voltage for this cycle), and charge from the memorycell capacitor is shared with the corresponding bit line andsubsequently with the internal sense amplifier node SA. From about t=22ns to t=23 ns, the voltage on the memory cell is decreasing while thevoltage on the bit line (not shown) and the internal sense amplifiernode SA is increasing. At around t=23 ns, about 80% of the availablesignal from the memory cell has developed on the sense amplifier nodeSA, and the bit line sense amplifier is strobed by a simultaneoushigh-going pulse on the sense enable signal SE and low-going pulse onthe complementary sense enable signal SEB. The simultaneous start timeof these two pulses (which is determined by an internally generatedtiming signal) occurs at a time interval “t₁” after the selected wordline is driven high, which is described in greater detail herein.

When the bit line sense amplifier is latched (at about t=23 ns), thehigh-going sense enable signal SE causes each of the selected bit linesense amplifiers to drive the lower of its two internal nodes (for thisexemplary cycle, internal node SAB) downward toward VSS for as long thesense enable signal SE is applied (e.g., pulsed) to the selected senseamplifiers. The fairly low capacitance of the internal sense amplifiernode SAB is then brought to VSS very quickly by one of the NMOStransistors in the bit line sense amplifier. The NMOS array selecttransistor (gated by the selected array select signal at VPP of about4.0 volts) then drives the low-going bit line (not shown) toward thecorresponding low-going internal bit line sense amplifier node SAB.Since the array select transistor is an NMOS transistor with a high gatebias above threshold, it has very low resistance and is able todischarge the low-going bit line relatively quickly compared to ahigh-going bit line (and memory, cell node). The sense enable signal SEis pulsed for a controlled time which is independent of thecomplementary sense enable signal SEB pulse time. Both pulses startsimultaneously, but each is terminated under independent control, as isadditionally described elsewhere herein. As seen in the figure, at aboutt=24 ns the sense enable signal SE returns quickly to VSS at a time, forexample, determined by several inverter delays after the end of the SEBpulse. Alternatively, the sense enable signal SE may remain high untilthe end of the active cycle, and then may be brought low at about thesame time as the selected word line is brought low (see, for example,alternative signal SE′ in FIG. 25).

The complementary sense enable signal SEB causes each of the selectedbit line sense amplifiers to drive the higher of its two internal nodesupward toward VDD for as long the complementary sense enable signal SEBfor the selected sense amplifiers is driven low. The array selecttransistor (gated by the selected array select signal now at VPP) thendrives the high-going bit line toward the corresponding high-goinginternal bit line sense amplifier node, and the memory cell accesstransistor (gated by the selected word line at VPP) then drives theinternal memory cell node toward the high-going bit line. If thecomplementary sense enable signal SEB were applied for a sufficientlylong time, then all three of these nodes would eventually charge inseries to substantially reach VDD, but this takes considerable time toasymptotically reach the final voltage. In contrast, as the waveformsshow, the complementary sense enable signal SEB is pulsed only for amuch shorter time (terminated by an internally generated timing signalST2 at about 24 ns), which leaves the internal sense amplifier node SAcharged to a voltage well below VDD, even though it was being chargedtoward VDD when the SEB pulse was active. Moreover, because of thedelays through the array select transistor, the bit line (not shown) ischarged to an even lower voltage than the internal sense amplifier node.Even more striking, because of the resistance of the bit line and thememory cell access transistor, the memory cell voltage has hardly beenrestored at all when the SEB pulse terminates. For example, at aboutt=23.6 ns the sense enable signal SEB returns quickly to VDD, and thevoltage on internal node SA stops rising with the memory cell node stillat only about 1.5 volts.

For the next approximately 2 ns (from about t=23.6 to t=25.4 ns), chargecontinues to be shared between the high voltage on the high-goinginternal sense amplifier node, the intermediate voltages along theresistive bit line, and the low voltage on the memory cell node becausethe selected array select signal and the selected word line both remainat VPP, and thus the array select transistor and the memory cell accesstransistor are relatively well turned on. Since the internal senseamplifier node and “near” end of the resistive bit line were chargedhigher than the desired final restored high level of approximately 2.2volts, they are discharged to a lower voltage by the charge sharing,while the far end of the resistive bit lines and the memory cell nodeare charged to a higher voltage by the charge sharing. The final voltageis reached more quickly than if the sense amplifier were fixed at thefinal voltage. In other words, once the distributed capacitance of thesense amplifier node, the resistive bit line, and the internal memorycell is decoupled from the charging current toward VDD (i.e., when thecomplementary sense enable signal SEB terminates and the PMOS senseamplifier transistor to VDD turns off), the “self-equilibrating” timefor these distributed nodes to reach approximately the same voltage issubstantially faster than if the memory cell and far end of theresistive bit line must rise to a fixed voltage (such as VDD) of thesense amplifier node and near end of the resistive bit line. Whensufficient charge has been shared between the high-going internal senseamplifier node, the high-going bit line, and the selected memory cell togenerate a predetermined restored high level on the selected memory cellof about 2.2 volts, this restored high level is carefully isolated inthe selected memory cell by bringing the selected word line back to VSS(followed, of course, by various equilibrating and precharging toprepare for the next active cycle), as is described in greater detailbelow.

The speed of this arrangement for restoring a predetermined high levelonto the high-going bit line and into the selected memory cell derivesfrom several separable factors. First, when enabled the PMOS transistorswithin the sense amplifier are driven toward VDD but stopped shortbefore the internal sense amplifier nodes fully reach the VDD level.This avoids a long exponential “tail” otherwise required to fully reachVDD. Second, the over-shoot of the sense amplifier node voltage, alongwith the self-equilibration of the distributed capacitance of the senseamplifier node, the high-going bit line, and the memory cell achieves afaster settling time on the selected memory cell (assuming a worst caseselected memory cell placement at the far end of the resistive bit line)to the desired high restore level. Third, the reduced voltage level of astored high is transferred by the array select and memory cell accesstransistors (both NMOS transistors with a fixed VPP-level gate voltage)with a lower time constant than if a higher voltage were transfer.

As is apparent from FIG. 25, the column select signal is driven to VDDsubstantially at the same time as the selected word line which, of note,is well ahead of signal developing on the internal sense amplifier nodesSA, SAB, and is brought back to VSS at about t=24.4 ns. The significanceof this timing will be described later in relation to FIG. 27.

Referring again to FIG. 25, an internal timing circuit generates atiming signal at the end of the timing interval “t₃” when sufficientcharge has been shared between the high-going internal sense amplifiernode, the high-going bit line, and the selected memory cell to generatea predetermined restored high level on the selected memory cell. Theactive cycle is then brought to a close by first bringing the selectedword line back to VSS, which occurs here at about t=25.4 ns. Immediatelythereafter (at about t=25.6 ns), the pulsed equilibrate signals (e.g.,SAEQ_PULSE, BLEQU, BLEQD, labeled as “PULSED EQ”) and the “level”equilibrate signals (e.g., SAEQ_LEVEL, not shown in FIG. 25) are drivenalmost simultaneously to VDD, which equilibrates the bit lines (at bothends) and the sense amplifiers to the bit line equilibrate voltage of,for this example, about 1.2 volts. The array select signals aresimultaneously returned to their inactive (VDD) level. The pulsedequilibrate signals are timed to automatically terminate when theequilibration accomplished by the pulsed signals is substantiallycomplete. The non-pulsed equilibrate signals (the “level” signals) stayactive until the next cycle (using this hole) is initiated.

The waveforms of FIG. 25 correspond to a particularly “fast”environmental corner (i.e., high VDD, cold temperature) and a typicalprocess having typical NMOS and PMOS transistors. Under theseconditions, there is a very short complementary sense enable signal SEB,but more importantly, there is a very fast rise time of the internalsense amplifier node SA which results in significant over-shoot of thehigh voltage level briefly achieved on the internal sense amplifier nodeSA before self-equilibrating with the distributed capacitance of theresistive bit line. In FIG. 26, the same cycle is shown at a “slow”environmental corner (i.e., low VDD, hot) with the same typicaltransistors. While all the waveforms behave generally as before (andtherefore merit little comment), some of the interesting (in some cases,subtle) differences will be pointed out.

A comment about performance is well taken at this point. As has beendescribed earlier, the various pulsed equilibrate signals (representedin the figure as “PULSED EQ”) are driven active at the end of a cyclefor a predetermined time, and automatically are brought back inactivewithout waiting for the next cycle to start, whereas the “level” (ornon-pulsed) versions of the equilibrate signals remain active until thestart of the next cycle and are then brought inactive early in thecycle. For a particular device operating at near its minimum cycle time,the level equilibrate signals may be brought inactive about the sametime as the pulsed equilibrate signals go inactive under automaticcontrol from the last cycle. Said differently, the “back sides” of boththe “level” equilibrate signals and the “pulsed” equilibrate signalssubstantially “line up” when a given device operates at its minimumcycle time. Looking again at FIG. 25, the unselected array select signalis brought to VSS at the start of the active cycle, which issubstantially the same time as the level equilibrate signal is alsobrought to VSS. Using the point when a signal crosses 1.4 volts as areasonable method to measure time (being approximately one-half of theVDD voltage), the level equilibrate signal therefore falls at a timelabeled 680. At the end of the cycle shown, the pulsed equilibratesignal falls at a time labeled 681, at which time the bit lineequilibration and bit line sense amplifier equilibration are complete.The falling edge of the level equilibrate signal in a subsequent cyclecould therefore line up with the falling edge of the pulsed equilibratesignal shown at point 681 for the current cycle. Measuring thehorizontal distance between points 680 and 681 on the horizontal axissuggests a minimum cycle time of 4.7 ns for the typical processembodiment shown operating at a fast environmental corner.

In FIG. 26, the VDD level is only 2.3 volts, but the VPP level remainsat approximately 4.0 volts because of its internal regulation withrespect to VSS, not VDD. This high, but regulated, VPP voltage helpsensure relatively low resistance of the “selected” array selecttransistors and memory cell access transistors without needlesslysacrificing reliability. The signal available from the memory cell isunfortunately less than before (due to the lower VDD). Consequently, the“t₁” timing interval is lengthened to allow more of the signal to bedeveloped on the sense amplifier nodes before sensing (i.e., the voltageof the memory cell is more completely discharged into the bit line/senseamplifier capacitance before strobing the bit line sense amplifiers).The “t₂” timing interval is also substantially lengthened to generate alonger pulse on the complementary sense enable signal SEB which isnecessary because the sense amplifier pull-up is much slower(asymptotically approaching a much lower VDD level relative to thedesired stored high voltage level, and at high temperature) and apredetermined amount of electronic charge (a “bucket of Q”) must beconducted into the sense amplifier/bit line nodes to generate the properhigh restore level. A consequence of this slower timing and slowertransistors is the lack of significant overshoot on the high-goinginternal sense amplifier node with respect to the bit line. That is,with the bit line sense amplifier node rising more slowly and being at alower voltage where the array select transistor is more conductive, thesense amplifier internal node voltage never exceeds the bit line voltagevery much. Therefore, less time is needed after the PMOS sensingterminates before the selected word line can be turned off. That is, the“t₃” timing interval is decreased.

FIG. 27 is a waveform diagram illustrating the major read path datasignals (for a non-burst cycle) for an exemplary low voltage, hightemperature read cycle, such as the cycle shown in FIG. 26. The internalsense amplifier nodes SA, SAB are shown on both figures to provide acommon reference. As has been described much earlier above, a selectedread amplifier (e.g., read amplifier 371 shown in FIG. 10) is enabled bya column select signal (and/or other similar signals depending on theparticular embodiment) to amplify the signal developing on the internalsense amplifier nodes and generate a level-shifted output signal on apair of local output lines LOUT, LOUTB, which are shared between a groupof eight sense amplifiers in a hole selected for read/write (e.g.,HOLE.W.X in FIG. 24). The amplified differential signal between LOUT,LOUTB is shown developing shortly after the SA, SAB signal begins todevelop. The large increase of the differential LOUT, LOUTB signal,which occurs at approximately t=24 ns, is a result of the bit line senseamplifier latching, and a substantially increased input signal beingamplified by the first stage local output amplifier. A multiplexer(within the local output sense amplifier block of, for example, FIG. 10)selects between a left pair and right pair of local output lines, and asecond stage amplifier then drives an associated pair of global outputlines GOUT, GOUTB, which extend vertically the full height of the memorybank, and provide the associated bit of data out either the top orbottom of the memory bank. The global output signals GOUT, GOUTB in FIG.27 are the delayed signals at the far end of the global output lines(which have distributed resistance and capacitance). At the top andbottom of the memory bank, for example, the data path continues with adifferential linear amplifier which is enabled to amplify and drive thedata signal from a selected pair of global output lines in a selectedmemory bank onto one end of a horizontally-arranged pair ofbi-directional, differential global data bus lines GDB, GDBB (waveformsnot shown), which route the read data signal to the physical outputbuffers (located, for this simulation, at the opposite end of the globaldata bus lines) whose first stage includes another differential linearamplifier which generates a pair of output nodes DOUT, DOUTB. Thedifferential voltage between DOUT and DOUTB is strobed by a latchingstage within the output buffer (e.g., latching amplifier 1083 in FIG.22B) which is the first stage in the data path which is powered by the“noisy” output buffer power supply terminals, rather than the more“quiet” internal power supply terminals which power the memory array andmost other internal supporting circuits. This latching buffer generatesa full rail-to-rail signal, which is then buffered by additional stageswithin the output buffer and ultimately driven off-chip. The latchingbuffer receives the input differential voltage on a pair of matched NMOStransistors. This differential voltage, near VDD, provides thedifferential current through these two NMOS transistors necessary tosteer the latching stage, even in the presence of about +/−1.5 volts ofnoise on either or both of the “noisy” power supplies with respect tothe voltages on the “quiet” supplies.

As can be seen from FIG. 27, there is about 60 mV of signal between DOUTand DOUTB, which is the signal at “the front porch” of the latchingoutput buffer, when the bit line sense amplifiers begin to be latched(at around t=23.7), and the magnitude of the signal grows substantiallyover the next 2.0 ns to a peak signal of about 400 mV (at t=25.5 ns).This latching output buffer stage is latched at “t₄” time (starting,perhaps, at a time between t=23.6 ns and t=25 ns), which is adjustableelectrically in a test mode and permanently with laser fuses based onthe results of an electrical test at wafer probe. As more imbalance oroffset exists in a particular data path, more signal is needed, beforestrobing a latch, to overcome the cumulative imbalances. One individualmemory device may have a worst case imbalance much smaller than anotherindividual memory device, and thus may function with smaller signal whenlatched. During final testing after assembly, the “t₄” time may betemporarily increased or decreased from it's permanently programmedvalue. Individual devices may be configured with as fast an access timeas possible, while still providing adequate signal margins in the outputbuffer latch. The t₄ timing for each device is preferably adjusted to beslightly later than the earliest t₄ timing for which the device stillfunctions properly, to ensure adequate margins when operating normally.A memory device can be final tested with the t₄ timing advanced relativeto its permanently programmed value set by the laser fuses, to ensuremargin at its less aggressive t₄ timing in normal operation.

FIG. 28 shows the same waveforms (except SA, SAB) but at a largervertical scale to more readily perceive certain small amplitude signals,and also shows the differential signal between GDB and GDBB, labeled as692, which was not shown in FIG. 27. For additional clarity, thedifferential signal between LOUT and LOUTB (labeled as 690), thedifferential signal between GOUT and GOUTB (labeled as 691), and thedifferential signal between DOUT and DOUTB (labeled as 693) are alsoindicated.

FIG. 29 is a waveform diagram illustrating the major array and senseamplifier signals when reading a low from the selected memory cell whileoperating at a VDD of 2.3 volts. It shows, of course, very similarwaveforms to those shown in FIG. 26, except for the selected memory celllevels. In FIG. 29, as one would expect when reading a low, the selectedmemory cell pulls the associated sense amplifier node downward (beforesensing) as charge is shared between the cell capacitor and the bitline/sense amplifier capacitance. Then, after sensing, the senseamplifier pulls the selected memory cell downward to relatively easilyrestore a low level of “substantially” VSS within the memory cell beforethe selected word line falls.

The next two figures highlight timing details of a memory cell andrelated circuitry during an internal write operation. FIG. 30 is awaveform diagram illustrating the major array and sense amplifiersignals when writing a high into the selected memory cell (having apreviously stored low) while operating at a VDD of 2.3 volts (i.e., aslow corner). In the embodiment whose waveforms are shown, the data fora write operation has already been driven onto the global input linesGIN, GINB in the form of a +/−100 mV signal by the write queue, allbefore the internal write operation begins. Then, during the actualcycle which carries out (or retires) the write operation, each pair ofglobal input lines (either all or a portion thereof corresponding to theparticular data bits to be written) is coupled to a selected bit linesense amplifier (e.g., by the circuitry of FIG. 11) when the columnselect signal is driven high, which occurs well before sensing. Thelarge capacitance of the (long) global input lines provides a readysource of electronic charge to drive the voltage of the sense amplifiernodes (as well as the bit lines connected thereto) to substantially thesame voltage as the global input lines, irrespective of the initialvoltage level of the selected memory cell which is also imparting chargeonto one of the bit lines and one side of the bit line sense amplifier.Thus, the high capacitance of the global input lines “swallows” most ofthe signal otherwise imparted by the selected memory cell, while theinitial differential voltage on the global input lines establishes thedesired differential write data signal onto the selected sense amplifiernodes. The bit line sense amplifier then latches according to the writedata signal coupled from the global input lines rather than latchingaccording to the previously stored data within the selected memory cell.

This action is clearly seen in FIG. 30. From about t=22.1 ns throught=23.6 ns, the selected memory cell is rising in voltage as its chargeis shared with, for this exemplary cycle, the true side of the bit linesense amplifier, internal node SA. But instead of decreasing in voltageas was seen in FIG. 29 when reading a low, here the voltage of internalnode SA is increasing due to the coupling of the true global input lineGIN (waveform not shown) to the internal sense amplifier node SA.1Moreover, on the non-selected-cell side of the bit line sense amplifier(the side coupled to the bit line which is not connected to the selectedmemory cell), the voltage of the complementary internal node SAB isdriven downward, rather than staying relatively unchanged as in the readcases above. The roughly 200 mV initial differential voltage which hadbeen developed between GIN and GINB prior to the beginning of the activecycle results in approximately a 175 mV signal between internal node SAand SAB just prior to sensing, even though the selected memory cell istrying to develop a nominal 100 mV signal of opposite polarity on theinternal nodes SA, SAB.

There is one major timing difference between read and write operationswhich is apparent from FIG. 30. In a write operation, the column selectsignal couples the low-capacitance sense amplifier nodes SA, SAB to thelarge-capacitance global input lines GIN, GINB, as described above,prior to sensing. However, the bit line sense amplifiers should be freeto latch, and drive one low capacitance internal node toward VDD and theother low capacitance internal node to VSS, without dragging the verylarge capacitance of the global input lines with it. Since thecomplementary sense enable signal SEB is a pulse timed to deliver arelatively predetermined amount of “Q” into the high-going senseamplifier nodes and bit lines, any undesired charging of any globalinput lines would rob some of this “packet” of charge and lower the highlevel restored onto the high bit line and thus into the selected memorycell. Consequently, in a write operation, the column select signal isbrought back to VSS (shown at about t=23.5 ns) just prior to thesimultaneous arrival of the true and complement sense enable signals SE,SEB. Then, the “restoration” of this just-latched high level into theselected memory cell proceeds identically as in a read operation. Notethat in a read cycle, the continued assertion of the column selectsignal after the simultaneous arrival of the true and complement senseenable signals SE, SEB provides continual development of thedifferential signal in the read path, but does not influence the bitline sensing operation, as is apparent from the read path 371 of FIG.10.

FIG. 31 illustrates a write operation at the same process corner whenwriting a low into a selected memory cell having a previously storedhigh. Of note, the selected memory cell, which was previously written toa high level of about 1.75 volts, is discharged into the true bit lineand true side of the sense amplifier, and yet the voltage of theinternal node SA is driven downward (and the voltage of thecomplementary internal node SAB is driven upward) by the coupling of theglobal input line pair to the sense amplifier. After latching, the senseamplifier restores a low level of approximately 50 mV into the selectedmemory cell, even though this cycle must fully discharge the selectedmemory cell capacitor from a stored high to a stored low.

As stated above, for a particular device operating at near its minimumcycle time, the level equilibrate signals may be brought inactive aboutthe same time as the pulsed equilibrate signals go inactive underautomatic control from the last cycle. Said differently, the “backsides” of both the “level” equilibrate signals and the “pulsed”equilibrate signals substantially “line up” when a given device operatesat its minimum cycle time. Looking again at FIG. 31, the unselectedarray select signal is brought to VSS at the start of the active cycle,which is substantially the same time as the level equilibrate signal isalso brought to VSS. Using the point when a signal crosses 1.2 volts asa reasonable method to measure time (being approximately one-half of theVDD voltage), the level equilibrate signal therefore falls at a timelabeled 720. At the end of the cycle shown, the pulsed equilibratesignal falls at a time labeled 721, at which time the bit lineequilibration and bit line sense amplifier equilibration are complete.The falling edge of the level equilibrate signal in a subsequent cyclecould therefore line up with the falling edge of the pulsed equilibratesignal shown at point 721 for the current cycle. Measuring thehorizontal distance between points 720 and 721 on the horizontal axissuggests a minimum cycle time of 5.3 ns for the typical processembodiment shown operating at a slow environmental corner.

Because the preferred embodiment uses only 128 word lines percomplementary bit line pair (64 memory cells connected to the true bitline BL, and 64 memory cells connected to the complement bit line BLB),and further because the preferred embodiment uses a first stage readamplifier connected directly to the internal nodes of each bit linesense amplifier, there is significantly less total capacitance on thecombined bit line/internal node than in a traditional design which uses256 word lines per complementary bit line pair. Even though the writtenhigh level is only approximately 2.0 volts, there nonetheless is moresignal available at the sense amplifier when sense enable occurs thanfor a traditional DRAM design using 256 word lines per bit line pair anda full VDD written/restored high level, even though not all of theavailable signal is used. For example, with aggressive timing, only 64%of the otherwise available signal (for extremely relaxed latch timing)may be actually achieved (e.g., 80% transferred to the bit line senseamplifier before sensing, and 80% transferred back into the selectedmemory cell before the end of restore), but this is still more signalthan for a traditional DRAM design using 256 word lines per bit linepair and a full VDD written/restored high level. Moreover, with shorterbit lines, especially when equilibrated from both ends, a much fasterequilibration time may be achieved.

FIG. 32 is a schematic diagram illustrating the preferred use of dualinput buffers for each address and control input for the memory arrayembodiment shown in FIG. 20, with one input buffer preferably locatedwithin the left spline, and the other input buffer located within theright spline. FIG. 32 further illustrates a timing compensation networkfor the internal clock signal which strobes the buffers, so that setupand hold times for both left and right buffers are closely matched. Aninput pad 731 for an address or control input is shown located on theleft side of the chip layout. An externally applied signal coupled tothe pad 731 is conveyed via a horizontally-arranged interconnect wire732 to a first input buffer 735 located in the left spline 565 and to asecond input buffer 738 located in the right spline 566. The horizontalinterconnect wire 732 is routed across the left-most memory bank 561(not shown) and across both central memory banks 562, 563 (not shown)through an otherwise unused wiring channel in one of the holes betweenarray blocks (see, for example, FIG. 20). A first R-C compensationcircuit 734 is provided between the interconnect wire 732 and thecorresponding “upstream” latching input buffer 735. This compensationcircuit 734 delays the input signal from reaching the upstream buffer735, and is sized to substantially match the additional delay of theinput signal in reaching the “downstream” buffer 738 which arisesbecause of the parasitic resistance 736 and capacitance of theinterconnect wire 732 traversing the width of two additional entirememory banks. If the arrival of the signal reaching the upstream buffer735 is delayed until the same signal reaches the downstream buffer 738,then both buffers 735, 738 may be clocked (i.e., strobed) at the sametime, and the apparent setup and hold time window required by thecombined use of two input buffers for the same input may besubstantially identical to that otherwise required if only one suchbuffer were used.

A second input pad 741 is also shown for a different address or controlinput located on the right side of the chip layout. An externallyapplied signal coupled to the pad 741 is conveyed via ahorizontally-arranged interconnect wire 742 to an upstream input buffer745 located in the right spline 566 and to a downstream input buffer 748located in the left spline 565. The horizontal interconnect wire 742 isrouted across the right-most memory bank 564 (not shown) and across bothcentral memory banks 563, 562 (not shown) through another otherwiseunused wiring channel in one of the holes between array blocks. An R-Ccompensation circuit 744 is provided between the interconnect wire 742and the corresponding upstream latching input buffer 745, which likewisedelays the input signal from reaching the upstream buffer 735 topreferably match the additional delay of the input signal in reachingthe “downstream” buffer 748 which arises because of the parasiticresistance 746 and capacitance of the interconnect wire 742 traversingthe width of two additional entire memory banks.

This interconnect wire 732 is preferably implemented as a metal wire,although other conductive materials might also be employed ifsufficiently low in resistance. The R-C compensation circuits 734, 744may be implemented as a lumped resistance and lumped capacitance, ormultiples thereof in series, but are preferably each implemented using adistributed resistance/capacitance structure to better match both thedelay and waveshape of the signal as received by the downstream buffer(which is delayed by the distributed parasitic resistance and thedistributed capacitance of its corresponding interconnect wire). Forexample, a long, narrow polysilicon feature, may provide adequately hightotal resistance, and may be loaded down with distributed capacitance toapproximately match the delay of the interconnect wire 732 between theleft and right splines.

A left internal clock signal CLK_L is conveyed on a vertically-arrangedwire 751 running up through the left spline 565, and strobes both theupstream input buffer 735 and the downstream input buffer 748. A rightinternal clock signal CLK_R is conveyed on a vertically-arranged wire752 running up through the right spline 566, and strobes both thedownstream input buffer 738 and the upstream input buffer 745. Both leftand right internal clock signals CLK_L, CLK_R are symmetrically drivenby a centrally-located clock driver 750 through a symmetricaldistribution network to substantially ensure phase-aligned clock signalsalong the full length of both the left and right clock signals.Consequently, a very short worst case setup and hold time is achievableover all such inputs (using an input buffer requiring valid data foronly a very short window of time). The use of a separate input buffer ineach spline for each input increases the input capacitance of each inputto the chip due to the long interconnect wire 732 or 742 (which inputcapacitance, of course, must be driven by the source of the externalsignal). However, each such input buffer now drives its complementaryinternal outputs only to decoder and control circuitry within the samespline. Thus, the total capacitive loading on the complementary outputsof each buffer are advantageously reduced. Furthermore, it easy to gateall the various signals to only the spline required for the particularoperation. For a read or write operation to memory bank 561 or 562, thevertical global control, timing, and address signals for spline 565 areactive, and the corresponding signals in spline 566 are inactive to saveconsiderable power. Likewise, when memory bank 563 or 564 is to be reador written, most of the signals in spline 565 remain inactive.

In an alternative embodiment, an “early” clock signal and a “late” clocksignal could be used in each spline. The upstream buffers within eachspline are then strobed using the “early” (i.e., “upstream”) clocksignals, and the downstream buffers are then strobed using the “late”(i.e., “downstream”) clock signals. In this case, the R-C compensationcircuits 734, 744 are not used, and the delay between the upstream anddownstream clocks is adjusted to substantially match the additionaldelay of the input signals, relative to their arrival at the upstreambuffers, in reaching the downstream buffers. In this way, the setup andhold time window may still be achieved (albeit at the additionalcomplexity of generating and distributing the second “downstream”clock).

FIG. 33 is a block diagram of an embodiment of a feedback controlledcircuit for generating an internal clock signal which is phase andfrequency locked to an external clock signal, and which is useful forstrobing address, data, and control input signals into the memory devicewith a setup and hold time window very closely aligned to and centeredabout the rising edge of the external clock.

An external clock signal EXT_CLK is conveyed on wire 761 to a voltagecontrolled delay line 763, which provides a delayed signal on its outputin accordance with an analog voltage received on a CTRL input (node765). The output of the voltage controlled delay line 763 is buffered bya buffer 764 to generate an internal clock signal CLK. The timing of theinternal clock signal CLK is thus delayed from the external clock signalEXT_CLK, and is adjusted by the voltage controlled delay line 763 tonominally be aligned with the next rising edge of the external clocksignal EXT_CLK, as is described in greater detail below.

The external clock signal EXT_CLK and the internal clock signal CLK areboth conveyed to a course adjust block 773 which functions to bring theinternal clock signal CLK in relatively close alignment to the externalclock signal EXT_CLK (by adjusting the voltage of the timing node 765).However, when the alignment is fairly close, the course adjust block 773then switches out, leaving just a fine adjustment block 776 to bring theinternal clock signal CLK to the final “close” alignment with theexternal clock signal EXT_CLK. Since the course adjust block 773 neednot provide for fine adjustment near the desired alignment, it may beadvantageously designed to quickly bring the internal clock signal CLKinto relatively close alignment with the external clock signal EXT_CLK.For example, during each clock cycle, it may cut the phase error byhalf, until the phase error is within +/−200 ps, at which point itswitches out, and provides no additional charge to or from the timingnode 765.

Within the fine adjustment block 776, the external clock signal EXT_CLKis also conveyed on wire 761 to a latching differential buffer 762,along with a reference voltage VRR which is preferably equal to one-halfVDD (for rail-to-rail external input signals, including the externalclock signal). The buffer 762 is strobed by the internal clock signalCLK. The internal clock signal CLK is delayed from the previous externalclock signal EXT_CLK, and the precise delay from the previous risingedge of the external clock signal EXT_CLK is fine adjusted by the fineadjustment block 776 to strobe the buffer 762 nominally at the nextrising edge of the external clock signal EXT_CLK. For example, assume agiven phase alignment of the internal clock signal CLK with respect tothe external clock signal EXT_CLK (which is already assumed to bebrought within +/−200 ps of the desired time). If the buffer 762, whenstrobed, interprets its input, EXT_CLK, as a logic “0,” (i.e., thecomplementary output is momentarily driven high, and the true outputremains low), then the external clock signal EXT_CLK during its risingedge must not yet have reached the reference voltage (e.g., one-halfVDD) when the buffer 762 was strobed. In other words, the buffer 762 wasstrobed too early.

If the complementary output of buffer 762 (node 768) is drivenmomentarily high, very narrow transistor 775 turns on very briefly toconduct a very small amount of charge from the timing node 765. Thetotal capacitance of the timing node 765 is the sum of both capacitor777 (the other terminal of which is coupled to VDD) and capacitor 778(the other terminal of which is coupled to VSS). Together, the totalcapacitance of the timing node 765 is relatively high. The small chargeremoved from the large capacitance of the timing node 765 results in avery slight decrease in voltage of the timing node 765, which is coupledto the CTRL input of the voltage controlled delay line 763. The voltagecontrolled delay line 763 then reacts to the lower voltage on its CTRLinput by very slightly adjusting its timing and delaying the phase ofthe internal clock signal CLK for the next (and following) cycles.

Conversely, if the buffer 762, when strobed, interprets its input,EXT_CLK, as a logic “1,” then the external clock signal. EXT_CLK duringits rising edge must have already exceeded the reference voltage (e.g.,one-half VDD) when the buffer 762 was strobed. In this case, the buffer762 was strobed too late. In this case, the true output of buffer 762(node 766) is driven momentarily high, and is inverted to drive the gateof P-channel transistor 774, which turns on briefly to conduct a smallamount of charge into the timing node 765, thereby increasing thevoltage on the timing node 765. The voltage controlled delay line 763then reacts to the higher voltage on the timing node 765 (coupled to itsCTRL input) by very slightly adjusting its timing and advancing thephase of the internal clock signal CLK for the following cycles.

In this fashion, the rising edge of the internal clock signal CLK isadjusted to nominally occur near the midpoint of the rising edge of theexternal clock signal EXT_CLK (i.e., be phase aligned with the externalclock signal). However, the internal clock signal CLK is actuallyadjusted to occur slightly ahead of the external clock signal EXT_CLK,because the circuit arrangement shown actually adjusts the setup andhold time window of the buffer 762, as strobed by the internal clocksignal CLK, to be centered around the mid-point of the rising edge ofthe external clock signal EXT_CLK. The fine adjustment block 776typically provides, during each cycle, a net change in charge on thetiming node 765, (and a resulting change of voltage on timing node 765)of a magnitude which causes about a +/−20 ps change in the delay throughthe voltage controlled delay line 763. The capacitance on the timingnode 765 is split between a first timing capacitor 777 connected to VDD,and a second timing capacitor 778 connected to VSS. The ratio of thesetwo timing capacitors is adjusted so that, for a change in voltage on,for example, VDD (as might occur during a noise spike), the voltage ofthe CTRL node is coupled by an amount which results in as little changeas possible in the delay through the voltage controlled delay line 763.

Additional buffers 771, 772, 767, 769 are implemented with identicalcharacteristics as buffer 762 (e.g., preferably using the same circuitand the same layout). For example, an external data input signal isreceived by buffer 772 which generates complementary internal datasignals DIN, DINB. The phase alignment of the internal clock signal CLKaffords a very narrow external setup and hold time window for theexternal data input signal which is substantially aligned to andcentered about the rising edge of the external clock signal EXT_CLK.Similarly, an external address input signal is received by buffer 771which generates complementary internal address signals Ax, AxB. Thephase alignment of the internal clock signal CLK likewise affords a verynarrow external setup and hold time window for the external addressinput signal which is substantially aligned to and centered about therising edge of the external clock signal EXT_CLK. To save power, thedata buffer 772 includes an enable input which is active only forexternal cycles which must strobe input data (e.g., those cyclesfollowing by one clock cycle (for the flow-through mode) or by twocycles (for the pipelines mode) the receipt of an earlier external writecycle, when the write data for such an external write cycle isreceived). Since the control inputs can initiate either a read or writeon any cycle, the control inputs and address inputs are strobed forevery external cycle. No such enable signal is provided, and the controlbuffers 767, 769 and the address buffers 771 are always enabled. WhileFIG. 33 illustrates the general relationship between the external clocksignal EXT_CLK, the internal clock signal CLK, and the strobing ofaddress signals by address buffers, the dual buffer arrangement of thepreceding figure is preferably employed in addition to the conceptsdescribed here in FIG. 33.

A first control signal CONTROL_1 (which might be, for example, aread/write control input) is conveyed to buffer 767, which samples thecontrol signal CONTROL_1 and conveys a corresponding output signal to acontrol circuit 770. A second control signal CONTROL_2 is conveyed tobuffer 769, which samples the control signal CONTROL_2 and also conveysa corresponding output signal to control circuit 770, which thengenerates internal control signals CTRL_A, CTRL_B which may variously becombinations of the external control signals CONTROL_1, CONTROL_2, andany other control signals CONTROL_N (not shown) and optionally, certainaddress inputs (as shown). Examples of such internal control signalsCTRL_A, CTRL_B include a signal to indicate a read operation from memorybank 1 (which requires the portion of the externally supplied readaddress that selects the memory bank to be read), a signal to indicate awrite operation to memory bank 2 (which requires, from the “bottom”entry of the write queue 779 (i.e., the oldest non-retired entry), theportion of the previously supplied write address that selects the memorybank to be written), a signal to indicate a refresh operation for memorybank 3 (which requires that memory bank 3 is neither performing a readnor a write), and other similar signals. By first sampling the externalcontrol signals in buffers 767, 769, then performing the necessarylogical operations on the output signals of the buffers 767, 769 togenerate the required internal control signals, the setup and hold timewindow for the control signals is well matched to that of the addressand data input signals (which is substantially aligned to and tightlycentered about the rising edge of the external clock signal EXT_CLK).

The internal clock signal CLK may also be used to control the turn-offtime of the data output buffers which drive the external data pins(e.g., and which are typically connected to an external data bus), sothat enabling another output buffer (to drive the same external databus) by a timing signal which occurs shortly thereafter (e.g., a timingsignal which occurs at one-third of the external clock period), does notresult in a bus conflict.

FIG. 34 is a layout diagram of a preferred embodiment of atwo-dimensional power supply bus grid within a memory bank for handlingthe very high transient current during bit line sensing. Tworepresentative array blocks ARRAY.X and ARRAY.Y within a memory bank areshown, along with adjacent holes HOLE.X.Y and HOLE.Y.Z. Looking first athole HOLE.X.Y, a VDD bus (labeled as 788) and a VSS bus (labeled as 789)are routed horizontally through the hole for the entire width of thememory bank. The VDD and VSS terminals of each bit line sense amplifierwithin the hole are respectively connected to the VDD bus 788 and theVSS bus 789 which run overhead each and every bit line sense amplifierwithin the hole.

However, the width of the VDD bus 788 and the VSS bus 789 is typicallylimited by layout constraints within the bit line sense amplifier. Ifthese are the only two buses providing a source of power and ground tothe entire row of bit line sense amplifiers within the hole HOLE.X.Y,the limited width of these buses results in a total bus resistance whichis high enough to significantly increase the time required to sense andrestore the high and low levels on the bit lines. Moreover, the bit linesense amplifiers near the lateral center of the memory bank would havedegraded voltages on their local portion of the power buses compared tothe sense amplifiers near the left and right sides of the memory bank(i.e., at the two ends of the VDD and VSS buses routed through thehole). Since the preferred embodiments of the present invention includesa complementary sense enable signal SEB which is a relatively shortpulse designed to deliver a predetermined amount of charge into thehigh-going sense amplifier and bit line nodes, such a wide variation oflocal power supply voltage would cause tremendous variations in thelevels written into various bit lines across the width of the memorybank.

To provide a much lower impedance VDD and VSS supply for each row ofsense amplifiers, a group of large VDD and VSS buses are provided whichare routed vertically (parallel to the bit lines) across the entireheight of the memory bank, and which respectively connect to each of thehorizontal VDD and VSS buses already connected to each sense amplifier,thus forming a two-dimensional grid of VDD and VSS buses. For example, avertical VDD bus 780 connects with the horizontal VDD bus 788 within thehole HOLE.X.Y (as indicated by the “dots” at their intersection), andfurther connects with the horizontal VDD bus 790 within the holeHOLE.Y.Z, and likewise connects with the horizontal VDD bus within allother holes (not shown). Similarly, a vertical VSS bus 781 connects withthe horizontal VSS bus 789 within the hole HOLE.X.Y, and furtherconnects with the horizontal VSS bus 791 within the hole HOLE.Y.Z, andlikewise connects with the horizontal VSS bus within all other holes(not shown).

As is described much earlier above (particularly in regard to FIGS.20-22), thirty-six differential pairs of global output lines (e.g.,GOUT, GOUTB) traverse the entire height of the memory bank and exit thetop of the memory bank to convey read data to an amplifier/multiplexer(which then conveys the selected half of these to associated data outputbuffers). Thirty-six additional differential pairs of global outputlines traverse the entire height of the memory bank and exit the bottomof the memory bank. Two differential pairs of these global output linesare grouped together, along with two differential pairs of correspondingglobal input lines, making a group of eight physical wires. Two sucheight-wire groups 782 and 785 are shown in FIG. 34. Thirty-six such8-wire groups are evenly spaced across the width of the memory bank(excluding a thirty-seventh 8-wire group for column redundancy) at aspacing (center-to-center) equal to approximately the width of sixteenbit, line sense amplifiers in one hole, or equal to approximately thewidth of 32 pairs of true and complement bit lines.

The vertical VDD bus 783 and the vertical VSS bus 784 together occupyalmost the entire gap between the adjacent wire groups 782 and 785.These vertical buses are preferably implemented in a high layer of metal(e.g., the top layer) and vertically pass directly above the bit lineswhich lie below (a few of the bait lines actually lie below the globalinput and output lines in wire groups 782, 785). Other VDD buses 780,786 and VSS buses 781, 787 are also shown, and which likewise occupy therespective gaps between their adjacent wire groups. These vertical VDDand VSS buses thus cover substantially all of each array block withinthe memory bank except for periodic groupings of global input and outputlines, which may be implemented in the same layer of metal as the VDDand VSS buses.

VDD-to-VSS filter capacitors are located at the top and bottom of eachmemory bank to provide substantial bypass capacitance to withstand thelarge current spikes which occur during sensing. These very widevertical VDD and VSS buses collectively provide a very low resistanceand very low inductance path between sense amplifiers located in eachhole and the VDD-to-VSS filter capacitors. The large metal buses allowthe stored charge in the filter capacitors to reach the two selectedrows of sense amplifiers with very little voltage drop, and allow theselected sense amplifiers to latch quickly and provide a good VSS lowlevel and uniform restored high levels to the bit lines within theselected array block. The filter capacitors, as well as other filtercapacitors implemented elsewhere within the device, are preferablyimplemented using many independent capacitors which are individuallyde-coupled and switched out of the circuit if more than a predeterminedleakage current is detected flowing through a given capacitor (i.e., a“shorted” capacitor).

Bit line crossover structures are advantageously used to achieve lowerworst case coupling, during both read or write operations, onto aparticular bit line pair from neighboring bit lines on either side. FIG.35, labeled prior art, is a layout diagram of a well-known bit linecrossover arrangement for reducing noise coupling from adjacent bitlines. A first complementary bit line pair BO, BOB is shownincorporating a lateral crossover at both the ¼ and ¾ points along itslength (which length, for the embodiments described herein, correspondsto the height of an array block). An adjacent complementary bit linepair B1, B1B incorporates a single crossover at the point half-way downits length. This pattern repeats every two pairs of bit lines, thus thethird bit line pair B2, B2B and fourth bit line pair B3, B3B areconfigured respectively like B0, B0B and B1, B1B. Becausephotolithographic guard cells are frequently used at the edges of eacharrayed group of memory cells, there is an increased layout area penaltyin providing crossover structures due to the required guard cells oneither side of each crossover structure. Eight such groups of guardcells, each labeled 800, are shown which are required with thisarrangement at the top and bottom edges of each arrayed group of memorycells. As indicated in FIG. 35, each guard cell group 800 may beimplemented as two additional non-functional (i.e., dummy) word lines.Consequently the area consumed by such guard cell groups is non-trivial,and thus the total area required to implement the crossover arrangementof FIG. 35 may be an appreciable percentage of the array block area.This prior art configuration reduces crosstalk (pattern sensitivity) andallows good signal development, but does so at the cost of significantextra area.

To reduce this area penalty, a novel crossover arrangement is employed,for certain embodiments, which provides a significant degree of noise(i.e., coupling) reduction, allows the same worst case signaldevelopment as the prior art structure, but requires only one crossoverstructure within each array block. Referring now to FIG. 36, eachcomplementary pair of bit lines runs vertically from the top to thebottom of an array block ARRAY.X, as before. The true bit line andcomplement bit line of a first pair (e.g., B1, B1B) run adjacent to eachother from the top to the bottom of the array block without anycrossovers. The true bit line and complement bit line of a second pair(e.g., B0, B0B) do not run adjacent to each other but instead straddlethe first pair, with a single crossover half-way down the second bitline pair (vertically in the middle of the array block). Both the trueand complement bit lines B1, B1B of the first pair lie between the trueand complement bit lines B0, B0B of the second pair. This crossoverarrangement repeats horizontally throughout each array block in groupsof two pairs of bit lines (four physical bit line wires). Thus, a thirdbit line pair B3, B3B and fifth bit line pair B5, B5B are configuredlike B1, B1B, and a fourth bit line pair B2, B2B and sixth bit line pairB4, B4B are configured like B0, B0B.

By using this crossover arrangement, only four groups of guard cells(each labeled as 801) are used in each array block—one each at the topand bottom of the array block, and one each at the top and bottom of thesingle crossover structure located in the vertical center of the arrayblock. Consequently the area consumed by such guard cell groups for thecrossovers as well as for the crossovers themselves is reduced to onlyone third that required by the prior art structure. And yet, the worstcase signal degradation due to capacitive coupling between neighboringbit lines is no worse than for the arrangement of FIG. 35. To moreeasily describe the coupling within the arrangement of FIG. 36, each bitline wire has also been labeled “A,” “B,” “C,” . . . “L.”

Consider first the coupling between wire B and C. Any change in voltagewhich develops on wire C (e.g., when accessing a memory cell connectedthereto) causes a change in voltage on wire B, albeit of a smallermagnitude, due to the lateral capacitive coupling between wires B and C.For example, if the voltage of wire C moves downward by 100 mV whenreading a memory cell having a stored low, the voltage of wire B maymove downward by, for example, 20 mV. The 100 mV of potential signalbetween the true and complement bit line (and likewise within the bitline sense amplifier 802) is reduced to a differential voltage of, forexample, 80 mV because of this self-coupling between adjacent true andcomplement bit lines. The presence of any crossover between the adjacenttrue and the complement bit line does not change this result, becausethe coupling therebetween remains unchanged. In other words, the priorart circuit suffers from this signal attenuation, also.

In contrast, the coupling between the B, C wire pair and the A, D wirepair is perfectly balanced, so there is no similar degradation ofdifferential signal levels on the B, C wire pair caused by voltageexcursions on either wire A or D, and likewise no degradation ofdifferential signal levels on the A, D wire pair caused by voltageexcursions on either wire B or C. For example, if the voltage of wire Cmoves downward by 100 mV, the voltage of wire D is coupled downward by acertain amount (e.g., 10 mV) due to the adjacent “C-D” coupling in theupper half of the array block, but the voltage of wire A is coupleddownward by the same amount due to the adjacent “C-A” coupling in thelower half of the array block. Thus, any differential voltage otherwisedeveloped on the A, D wire pair is unaffected by any voltageperturbation of either wire B or C. (As the example just describedshows, however, the common-mode voltage levels of the A, D wire pair maybe affected by voltage perturbations of either wire B or C.) Similarly,if the voltage of wire A moves downward by 100 mV, the voltage of wiresB and C are coupled downward by the same amount (e.g., 10 mV), and thedifferential voltage otherwise developed on the B, C wire pair isunaffected by any voltage perturbation of either wire A or D.

An arbitrary word line WL is shown in the upper half of the array block,and the memory cells driven by the word line WL are connected toparticular bit lines as indicated by the open circles. Assume that eachof these memory cells store the same data, which is a low, when the wordline WL is driven high. The voltage of each bit line consequently movesin a direction as indicated by the solid vertical arrows. Wire H iscoupled downward (i.e., “the voltage of wire H is coupled downward”) bythe “I-H” coupling in the upper half of the array block (represented bya small lateral capacitor) and by the “A-H” coupling in the lower halfof the array block (also represented by a small lateral capacitor). WireH is thus coupled downward (indicated by a dashed arrow) along itsentire length by “an adjacent bit line wire” (actually half of wire Iand half of wire A), which is no worse than the self-coupling betweenwires B and C described above or any pair of adjacent (crossing) wiresof the prior art structure. Note that for this arbitrary word line WL,wire D is not driven by a memory cell and therefore does not couple asignal to the top half of wire E. Similarly, wire L does not couple asignal to the bottom half of wire E. Therefore, the E, H wire pair is noworse than the B, C wire pair. That is, for the signals and capacitancesof the example, the prior art arrangement of FIG. 35 provides a signalof 80 mV independent of what the neighboring bit lines do. The newarrangement of FIG. 36 provides a worst case signal of 80 mV, no worsethan the prior art Interestingly, it provides a best case signal of 120mV (wires A and I having opposite data as wire E) which is of littlevalue since the memory device must function under worst case conditionsor patterns. An analogous situation arises for a data pattern of allhighs stored within each memory cell driven by a given word line.Consequently, the worst case pattern sensitivity of this arrangement isan array of all 1's or all 0's, which is particularly helpful inreducing required test times.

The crossover structure shown in the embodiment thus far described islocated at the mid-point of the respective bit lines (i.e., half-wayacross the array block). As described, this cancels the non-common-modecontribution of the lateral capacitive coupling from neighboring lineswithin the array block. For some embodiments, a location other than themid-point may be preferred. For example, the lateral coupling betweenadjacent bit lines which occurs within a portion of the bit line senseamplifier layout, or within a portion of an equilibration circuit, orfor some other reason, may contribute to a coupling onto one of a pairof bit lines which is not entirely offset by an equal coupling onto theother of the pair of bit lines when the crossover structure is locatedat the mid-point of the array block. As a result, the placement of thecrossover structure may be at a location within the array block otherthan half-way across the array block to provide a substantially equalcoupling onto both bit lines of the pair (i.e., substantially negligibledifferential coupling).

It should be appreciated that a crossover structure as used hereinprovides an electrical path for one line to cross another line withoutcontacting the other line. When viewed in layout terms, it does notnecessarily imply that a higher level interconnect layer must be used to“cross over” a lower level interconnect layer. In other words, a “firstwire which crosses over a second wire” may be used herein withoutseparate meaning from a “first wire which crosses under a second wire”unless the context clearly requires a distinction. It shouldconsequently be appreciated that a crossover structure which provides apath for a first wire to “cross over” a second wire may be implementedusing either a lower or higher interconnect layer than the second wire.Moreover, while described in the context of a dynamic memory array, thecross-over arrangement shown in FIG. 36 is also well suited for othertypes of arrays whose memory cells connect to only one of either a truebit line or a complement bit line, such as a read only memory array or aprogrammable read-only memory array. The arrangement of FIG. 36 is notas well suited for use with static memory arrays because the spacingbetween a given true and complement bit line (at least for half of thebit line pairs) makes connection of a static memory cell to both thetrue and complement bit lines somewhat more difficult.

FIG. 37 is a timing diagram for several interspersed external read andwrite cycles for an exemplary embodiment of the invention. The diagramillustrates the timing of external address and data signals, and severalimportant internal data signals, when operating in the pipelined mode ofoperation (rather than the flow-through mode of operation). The upperwaveform depicts the external clock signal and is labeled to show ninesuccessive cycles T1, T2, . . . T9, each initiated by the correspondingrising edge of the external clock signal. The second waveform representsa read/write control signal R/W# which is strobed, along with theexternal address and data signals, at a time generally corresponding tothe rising edge of the external clock signal (as described above). Ifthe read/write control signal R/W# is high when strobed, the cycle thusinitiated is a read cycle, and if the read/write control signal R/W# islow when strobed, the cycle thus initiated is a write cycle. For ease ofreference, each cycle T1, T2, . . . T9 is also labeled as either “R” or“W” accordingly to more easily identify each cycle as either a read orwrite cycle.

The third and fourth waveforms illustrate the external address and datasignals corresponding to each cycle, which are conveyed respectively onthe external address and data pins of the memory device. An externaladdress signal (i.e., an N-bit address) is strobed into the memorydevice during each cycle irrespective of whether the cycle is a read orwrite (or idle) cycle. The external address signal presented to andstrobed into the memory device for the T1 cycle is labeled A1, theexternal address signal strobed into the memory device for the T2 cycleis labeled A2, and so forth, for all nine cycles shown.

Since the embodiment described in FIG. 37 incorporates a common data I/Obus, the external data signals are indicated as “DOUT(address)” toindicate the memory device is driving the external data pins with a readdata signal corresponding to the “address,” or are indicated as“DIN(address)” to indicate the memory device is receiving a write datasignal on the external data pins corresponding to the “address.” Forexample, the external data signal indicated as DOUT(A2) is the read datasignal corresponding to the address A2 which is driven by the memorydevice onto its external data pins, while the external data signalindicated as DIN(A3) is the write data signal corresponding to theaddress A3 which is presented to the external data pins and received bythe memory device.

In the exemplary embodiment when operating in the pipelined mode ofoperation, the external address for a read cycle is strobed by a firstrising edge of the external clock (e.g., the T2 rising edge), and thecorresponding data which is read from the selected memory cells isdriven onto the external data pins after a second rising edge of theexternal clock (e.g., the T3 rising edge), to be valid at the circuitreceiving the data at a third rising edge of the external clock (e.g.,the T4 rising edge). If an external write cycle follows immediatelyafter two external read cycles, the write address is presented to thememory device on its external address pins and strobed into the devicejust like for a read cycle (on the rising edge of the external clock),but the external data pins are occupied with driving the read datasignal corresponding to the first external read cycle and cannot be usedat this time to present the corresponding write data signal. In the nextcycle, the external data pins are occupied with driving the read datasignal corresponding to the second external read cycle, and again cannotbe used to present the corresponding write data signals. Instead, thewrite data for the external write cycle is driven onto the data bus andpresented to the device during the cycle in which output data would haveappeared had the cycle been an external read cycle instead of anexternal write cycle. In other words, in a write cycle the write commandand the write address are concurrently presented to the memory deviceand strobed by a rising edge of the external clock, but thecorresponding write data is presented to and strobed into the device bythe rising edge of the external clock which occurs two cycles later. Inthis way, the address bus (i.e., the external address pins) and the databus (i.e., the external data pins) are used every cycle, with no wastedcycles for either bus. The address and data for the write cycle are bothstored into an internal write queue, and the actual internal writeoperation to physically store the write data into the selected memorycells is postponed until some subsequent write cycle.

Read bypass circuitry is provided which allows data corresponding to theaddress of the read cycle to be correctly read from the write queuewhenever an earlier queued write directed to that same address has notyet been retired (i.e., written into the appropriate memory cells). Whena read cycle is initiated, it immediately starts to access the addressedlocations within the memory arrays, and simultaneously compares its readaddress to all the pending addresses in the write queue. If an address“hit” occurs to an entry in the write queue, then the data from allbytes that were enabled during the write (for that entry) are obtainedfrom the write queue rather than from the memory array. If more than one“hit” occurs, the most recently written data (on a byte-by-byte basis)is substituted for the data from the memory array, and any bytes whichwere not enabled for writing by any entry in the write queue areretrieved from the memory array. Thus, data not yet retired from thewrite queue can still be properly read during a subsequent read cycle.

A representative read cycle's internal timing may be appreciated byexamining the T2 cycle (which is assumed to be a non-burst cycle). Asstated above, the rising edge of the external clock signal which startsthe T2 cycle strobes the corresponding address signal A2 into the memorydevice. The control signals, including a READ/WRITE# control signal, arealso received and the cycle determined to be a read cycle. An internalread operation is initiated which uses the strobed external addresses,decodes a selected word line in a selected array block of a selectedmemory bank, enables the bit line sense amplifiers for sense/restore inthe holes both above and below the selected array block, and likewisedecodes a selected column select signal and other related read signals(e.g., READ_L or READ_R, READ) within one of the holes adjacent to theselected array block (either the hole above or below the selected arrayblock), all as has been extensively described above. The respectivesignals from the thirty-six selected sense amplifiers are amplified anddriven onto thirty-six corresponding vertical pairs of global outputlines GOUT, GOUTB (eighteen of which exit the memory bank at the top,and the remaining eighteen of which exit the memory bank at the bottom).The fifth waveform indicates the development of this signal on theselected group of global output lines, labeled representatively as GOUT.A read signal which results from the read data at the A2 address(indicated by an arrow labeled 840) develops on the global output linesduring a later portion of the T2 cycle, and consequently is labeled R2.Recall that each true and complement global output line is preferablyloaded by a static “resistive” load device (a grounded-gate PMOStransistor) to VDD. When enabled, each local output amplifier driving acomplementary pair of global output lines pulls one line of the pairdown (e.g., the true global output line) while leaving the other line atVDD to generate a differential read signal on the complementary pair ofglobal output lines. The voltage of both true and complement globaloutput lines GOUT, GOUTB is driven by the load device back to VDD (or,if already at VDD, is held at VDD) when no local output amplifier isenabled to drive the complementary pair of global output lines. Thisoccurs automatically near the end of a cycle when the column selectsignal and associated read signals are de-activated, which is describedin greater detail elsewhere herein, particularly in regard to FIGS. 8and 10. The global output waveforms are drawn here in a manner to remindthe reader of the general voltage levels and timing of these globaloutput lines.

The respective signals from the thirty-six vertical differential pairsof global output lines GOUT, GOUTB are amplified and driven ontothirty-six corresponding horizontal differential pairs of bi-directionalglobal data bus lines GDB, GDBB (eighteen pairs of which runsubstantially generally across the top of the memory array, and theremaining eighteen pairs run substantially generally across the bottomof the memory array). The sixth waveform indicates the differentialsignals present on these global data bus lines, and is representativelylabeled GDB. The read signal on the selected global output lines duringT2 is buffered to generate an associated read signal on the global databus lines (indicated by an arrow labeled 841) which also corresponds tothe read data at the A2 address. Consequently, the global data bussignal which develops during T2 is also labeled “R2”. Since each trueand complement global data bus line is also loaded by a static“resistive” load device to VDD, the behavior of the global data bus isvery similar to that of the global output lines. As such, the globaldata bus waveforms are drawn here in a similar manner to remind thereader of the general voltage levels and timing of these global data buslines.

The read signal on each of the global data bus pairs is furtheramplified by yet another linear amplifier (which is preferably locatednext to the output buffer) and subsequently strobed into a registerwithin each respective output buffer, all within the T2 cycle (orperhaps, in the case of a memory device operating with a very shortcycle time, near the beginning of the next cycle). However, the data outsignal is not driven onto the external data pins during the current T2cycle, but rather during the next (T3) cycle (for this embodiment,operating in the pipelined mode of operation). The arrival of the nextrising edge of the external clock signal which initiates the T3 cyclealso causes the output buffer to drive onto the external data pins(indicated by arrow 847) at a predetermined time within the T3 cycle(e.g., ⅓ of the way into the cycle), the data signal associated with theprevious read operation performed during the T2 cycle, which data signal(R2) is latched from the global data bus, as indicated by arrow 842.Thus, the data signal driven onto the external data pins during the T3cycle is labeled DOUT(A2).

This next (T3) cycle happens to a write cycle (for the exemplary cyclesshown). The address A3 is strobed as in the read case, but no internalwrite operation of the memory array is initiated for the A3 address,because the data to be written is not yet available within the memorydevice Instead, an internal operation of the memory array is initiatedfor the another write address (which, for example, could be the Wxaddress already stored within and now at the bottom of the first-in,first-out write queue), and the A3 write address is stored into the topof the write queue. The write data for the T3 cycle is presented laterto the memory device on the external data pins during the time that theexternal data pins would otherwise have been driven, if the T3 cyclewere instead a read cycle, with read data corresponding to the A3address, namely during the T4 cycle (more specifically, from a timesomewhat after the T4 cycle begins until a time somewhat after the T5cycle begins). As is indicated by arrow 848, the T4 rising edge quicklycauses the data output buffers to cease driving the DOUT(A2) data signalonto the external data pins. Soon thereafter, another device providesthe data to be written at the A3 address, DIN(A3), to the external datapins, still during the T4 cycle, and held valid into the T5 cycle, sothat the rising edge of the external clock signal (i.e., the T5 risingedge) can strobe the write data presented to the external data pins,which is then also stored into the write queue with the previouslyreceived and stored A3 write address.

The table at the bottom of FIG. 37 indicates, for cycles T3 through T8,whether an internal array operation is carried out (and if so, whetherread or write), the number of holes which are enabled for bit line senseamplifier sense/restore (whether 0 or 2), the number of holes which areenabled for write (whether 0, 1, or 2), and the number of bits writtenby an internal operation during the cycle (whether a 36-bit single wordor a 72-bit double word) and the particular address associated with thedata written (i.e., the “identity” of the data written).

During all external write cycles, the write queue drives thebi-directional global data bus GDB with a write data signalcorresponding to an earlier external write cycle. As a result of thespecific order of the exemplary read and write cycles described in FIG.37, the write queue, during the T5 cycle, drives the bi-directionalglobal data bus with a write data signal corresponding to the data to bewritten at address A3, which is labeled as W3. This causal relationshipis indicated by an arrow labeled 843. (Other cycles described belowshould make it clear that for some cycles, the internal write datasignal is not necessarily driven onto the global data bus during thesame cycle which strobes the write data.) In the exemplary embodiment,recall that the internal data path into each memory bank is twice aswide (i.e., a “72-bit double word”) as the external I/O word width(i.e., the least significant address bit selects either a lower 36-bitword or an upper 36-bit word). Once a write data signal is placed ontothe global data bus, a corresponding signal is driven onto thethirty-six complementary pairs of global input lines GIN, GINB which areselected by the write address A3 (specifically, the two bits of addressthat select one of the four memory banks, and the least significantaddress bit that selects between the upper and lower 36-bit data words)previously stored into the write queue. As indicated by the arrowlabeled 844, the corresponding W3 write data signal is driven onto, forthis exemplary cycle, the group of thirty-six pairs of global inputlines associated with the lower 36-bit word in the selected memory bank,which is labeled GIN(LW).

The voltage levels and timing of the global input lines are brieflyreiterated here to further clarify the cycle-to-cycle timingrelationships shown in FIG. 37. An internal write operation is performedduring each cycle which is initiated with the read/write control signalR/W# at a low level—namely during each external write cycle (except fora merged write, described below). However, the internal write operationperformed during a given external write cycle corresponds to a writeaddress and write data previously strobed into the device and storedwithin the write queue. At the end of each such internal writeoperation, the thirty-six complementary pairs of global input lineswhich are addressed by the next write operation are driven with therespective write data signal appropriate for the next internal writeoperation (which are also already stored within the write queue). Noneof the remaining complementary pairs of global input lines change state.Since, for the preferred embodiment, there is both a first group of 36complementary pairs of global input lines associated with the upper36-bit word (i.e., the “upper-word global input lines”) and a secondgroup of 36 complementary pairs of global input lines associated withthe lower 36-bit word (i.e., the “lower-word global input lines”)foreach of the four memory banks, there are eight total groups of globalinput lines, each group having 36 complementary pairs of global inputlines. In other words, at the end of a given internal write operation,one particular group of thirty-six complementary pairs of global inputlines (which are addressed by the next write operation) are driven withthe appropriate write data (for the next internal write operation), andthe other seven groups remain unchanged.

During the T5 cycle (which is an external write cycle), a pending writeis retired from the write queue by performing an internal writeoperation which writes a data signal Wz into the 36-bit word addressedby a corresponding address Az, both of which were presented to thememory device and strobed into the write queue earlier than any of theexemplary cycles shown here. The 36-bit word decoded by the address Azis the lower of the two words, and thus the write queue (at the end ofthe T4 cycle) drives the data signal Wz onto the lower-word global inputlines GIN(LW) in preparation for the early part of the T5 cycle, whenthe data signal Wz is utilized (because, like any other write cycle, theT5 cycle expects to retire the bottom write queue entry, unless a mergedwrite).

The internal write operation is accomplished, during the early portionof the T5 cycle, by coupling each of the global input line pairs to thecorresponding bit line sense amplifier which is selected by the decodedwrite address from the write queue, prior to sensing the bit line senseamplifier. The signal otherwise developed by the selected memory cellsis swallowed by the larger capacitance of the global input lines, andthe bit line sense amplifiers latch according to the write data signaland then restore the bit lines voltages accordingly. This functionalityis described in great detail elsewhere herein. At the time the bit linesense amplifiers begin to sense, the internal nodes of the bit linesense amplifier are decoupled from the global input lines. Consequently,near the end of the internal write operation, the data signals for thenext internal write operation are driven onto the particular group ofglobal input lines which are addressed by the next internal writeoperation. This is performed by first equilibrating the prior datasignal from the next group global input lines, then by driving each trueand complement global input line with a controlled current for acontrolled amount of time, resulting in the high-going global input linemoving up in voltage by about 100 mV, and the low-going global inputline moving down in voltage by about 100 mV. As described earlier, thissignal is developed on each corresponding pair of global input linesbefore the equilibration of the bit lines is complete. As shown in theFIG. 37, the new data signal W3 is developed on the lower-word globalinput lines well in time for the next cycle, T6, to use this data signalprior to bit line sensing, should that cycle have been a write cycle.

However, since the T6 cycle is a read cycle, an internal read operationis performed rather than an internal write operation. Consequently, thedata signal W3 remains dynamically floating on the particular group of36 global input lines until the next external write cycle, in this casethe T7 cycle. At this point in time the write operation to address A3 isthe oldest entry (i.e., not yet retired) in the write queue. But thewrite operation to address A4 has also been written into the queue alongwith the corresponding data signal DIN(A4). During the T7 (write) cycle,this data signal is driven onto the bi-directional global data bus(indicated by the arrow labeled 845 writing the data signal labeled W4).In the exemplary cycles shown, the A4 address is assumed to be equal tothe A3 address but with the LSB complemented (e.g., when A4=A3+1 orA4=A3−1).

The selected memory cells corresponding to two addresses which differ inonly the least significant address bit (LSB) correspond to the same72-bit double word, as described above. Before an internal writeoperation is performed, the address of the pending write operation iscompared to the address of the next write operation (which has alreadybeen presented to the memory device during an earlier external writecycle and is already stored in the write queue). If the selected memorycells to be written in both the pending (first) write operation and inthe next (second) write operation correspond to the same 72-bit doubleword (i.e., differ in only the least significant address bit), thepending internal write operation which would otherwise follow from thefirst external write cycle is not yet carried out. In the exemplarycycles shown, the data signal W3 remains on the lower-word global inputlines (for the memory bank decoded by both address A3 and A4) while thedata signal W4 is generated on the upper-word global input lines duringthe T7 cycle and both data signals remain dynamically on the respectiveglobal input lines until the next external write cycle, which in thiscase is the T8 cycle.

During the T8 cycle, the address of the double-word pending writeoperation is again compared to the address of the next write operation,A5, which for this example is assumed to decode into another memorybank. In other words, the selected memory cells to be written in boththe double-word pending write operation and in the next write operationdo not correspond to the same address (ignoring, of course, the LSB).Consequently, a single internal write operation is carried out duringthe T8 (write) cycle which simultaneously writes both the 36-bit dataworld initiated by the T3 external write cycle and the 36-bit data wordinitiated by the T4 external write cycle. Recall that in any cycle, thehole above the selected array block and the hole below the selectedarray block are enabled for sense/restore. In a write cycle which writeseither the upper or lower 36-bit word, one of these two holes is alsoenabled for write (see FIGS. 8, 11, 21, and 24, and relateddescription). To write both the upper and lower 36-bit words, such asoccurs in a merged write cycle, both holes above and below the selectedarray select are also enabled for write. At the end of the T8 cycle,both the lower-word global input lines GIN(LW) and upper-word globalinput lines GIN(UW) remain unchanged—neither is equilibrated and drivento a new data signal. This results because, for the exemplary cyclesshown, the next address in the write queue (A5) decodes to select adifferent one of the four memory banks, and consequently the particulargroup of global input lines which is driven in preparation for the nextwrite operation is associated with the memory bank to be written in thenext write cycle.

By merging external write cycles having sequential addresses into oneinternal write operation, a significant amount of internal powerconsumption is saved compared to performing two separate writeoperations since the selected memory bank is cycled only once (insteadof twice) to write the two words. An external write cycle is alwayscarried out (i.e., retired) exactly three write cycles later unless itis delayed by the write merging with the following write cycle. Lookingagain at the relative timing of the particular cycles shown, theexternal T3 write cycle is retired four external write cycles later, dueto the write merging. The normal latency of three write cycles resultsfrom several factors: the bi-directional external data bus (i.e.,external date pins) which is operated with no wasted cycles in a“zero-bus-turnaround” manner (in accordance with the ZBT® protocol, aregistered trademark owned by Integrated Device Technology, Inc.), thebi-directional global data bus GDB (also operated with no wasted cyclesin a zero-bus-turnaround manner), the timing of driving the global inputlines GIN late in a cycle (to avoid driving them during bit linesensing) and the method used to accomplish the write during the earlyportion of a cycle by swallowing the read signal otherwise developed inthe bit line sense amplifier. Two external cycles are required beforethe external data signal for an external write cycle is strobed andavailable within the memory device and stored within the write queue.Only during an external write cycle is this data driven onto thebi-directional global data bus GDB, which may occur during the samecycle which strobes the data if the external cycle happens to be anotherexternal write cycle (such as occurs during the T5 cycle), or may occurduring some subsequent external write cycle if the external cycle whichstrobes the data happens to be an external read cycle (such as occursduring the T6 cycle).

During the particular cycle that a particular write data signal isdriven onto the global data bus, it is also buffered and driven onto theselected group of global input lines near the end of the particularcycle in preparation for the next internal write operation. The actualinternal write operation is then performed (unless it is delayed andmerged with the following internal write operation) during the nextexternal write cycle.

Write cycle merging has been described thus far in light of two writecycles, each writing a full 36-bit word, into two sequential memoryaddresses (differing only in LSB). Write cycle merging is also usefulwhen sequential external write cycles write different portions of thesame 36-bit word, or when sequential external write cycles over-writesome or all of the same 36-bit word, using the byte write capability ofthe preferred embodiment. In each of these cases, like the othersdescribed above, the selected memory cells to be written in both thepending write operation and in the next write operation correspond tothe same 72-bit double word (i.e., having addresses which differ only inLSB), and the pending internal write operation which would otherwisefollow from the first external write cycle is not carried out but ratheris merged with the next internal write operation. As an example, foursequential external write cycles, each writing a different (orover-writing the same) 9-bit byte within a 36-bit word corresponding toa given address, followed by four more sequential external write cycles,each writing a different (or over-writing the same) 9-bit byte within a36-bit word at an address which differs from the given address only inthe LSB, is actually carried out internally as a single internal writeoperation, simultaneously writing all 72-bits (assuming all 8 bytes werebyte-write enabled in at least one of the eight cycles) into theselected memory cells. In the event the same 36-bit single word (or aportion thereof) were written in two consecutive external write cycles(both preceded and followed by write cycles to addresses whichcorrespond to a different 72-bit double word than the two write cyclesin question), the two write cycles would merge, but only one hole (aboveor below the selected array block, but not both) would be enabled forwriting. That is, a hole is enabled for writing by the need to writedata into that hole, not just by the fact that two write cycles havebeen merged.

It should be noted that READ cycles could be merged at the expense of anaccess time penalty, but there is NO such penalty for merging writecycles. To merge read cycles, all internal read operations would be72-bits wide, and a comparison of the read address (just received forthe current cycle) to the previous read address would be required beforedeciding whether to start an internal read operation, or whether theread data is already available as a result of the last read operation.

The ZBT® protocol (in which the write data for the external write cycleis driven onto the data bus and presented to the device during the cyclein which output data would have appeared had the cycle been an externalread cycle instead of an external write cycle) requires a write queue.In conventional writing, data is written into the memory array on thecycle the data and its address are received. But even for moreconventional write timing, including the case where write data isreceived at the same time as the write address and write command,delaying the write with a write queue offers several advantages overconventional writing. First, it allows the global input voltages to bechanged at a time in the cycle that will not disturb the sensing of thebit lines or sense amplifier nodes below. Second, it allows merging ofwrite cycles, since the next address to be written already exists on thechip. Third, it allows a write cycle to be aborted in the event there isa (parity) error in the address field during a write command. If notaborted, such an error would overwrite data at some address, the data atthat address lost forever. The ability to abort a write cycle upon anerror could be done without a write queue, but would delay the writeuntil a parity check could be performed.

FIG. 38 is a schematic diagram of a latch timing circuit 930 forgenerating the major timing signals which control the sense and restoretiming of the bit line sense amplifiers. This latch timing circuit 930is used to time the simultaneous start of both NMOS and PMOS sensingrelative to the timing of the selected word line being driven high, totime the end of PMOS sensing, and to time the simultaneous end of NMOSsensing and the selected word line being brought low (which isimmediately followed by bit line and bit line sense amplifierequilibration).

In a preferred embodiment, the latch timing circuit 930 produces threemain timing signals, ST1, ST2, and ST3. The first timing signal, ST1, isused to control, relative to the timing of the selected word line beingdriven high, the simultaneous start of both the NMOS and PMOS sensing.The second timing signal, ST2, is used to control, relative to thesimultaneous start of NMOS and PMOS sensing, the duration of the PMOSsensing, and the third timing signal, ST3, is used to control, relativeto the end of the PMOS sensing, when to simultaneously end the NMOSsensing and bring the selected word line back low. Each of these timingsignals ST1, ST2, and ST3 is independently adjustable (described indetail below) and respectively defines a corresponding timing intervalt₁, t₂, and t₃. The timing interval “t₁” begins with the selected wordline being driven high and ends with the simultaneously start of boththe NMOS and PMOS sensing (i.e., the timing interval “t₁” is the amountof time the selected word line is high before sensing). The timinginterval “t₂” extends from the simultaneous start of NMOS and PMOSsensing to the end of PMOS sensing (i.e., the timing interval “t₂” isthe duration of the PMOS sensing). The timing interval “t₃” extends fromthe end of the PMOS sensing to the simultaneous end of the NMOS sensingand discharge of the selected word line (i.e., the timing interval “t₃”is the amount of time the word line remains high after the end of PMOSsensing). The adjustment of each of these timing intervals t₁, t₂, andt₃ is made possible by the adjustment of the three respective timingsignals ST1, ST2, and ST3, and is described in greater detail below.

The timing interval t₁ essentially controls how much signal from thememory cell reaches the sense amplifier before starting the NMOS andPMOS sensing. A short t₁ does not provide enough time for all the chargein a selected memory cell (especially one located at the “far end” ofthe resistive bit lines, furthest from its bit line sense amplifier) tofully share with the charge on the bit line and sense amplifier nodes,and consequently the sense amplifier begins to sense with less signalthan would be developed if, alternatively, a longer t₁ were configured.A longer t₁ increases operating margins (i.e., bit line sense amplifierdifferential voltage) at the expense of increased cycle time.

Similarly, the timing interval t₂ essentially controls how much chargeis driven onto the high-going sense amplifier node, bit line, and(high-going) memory cell during sensing. Increasing t₂ increases thevoltage stored into the memory cell, but also increases the bit lineequilibrate voltage when charge is later shared between true andcomplement bit lines (and sense amplifier nodes). A short t₂ may notprovide enough charge to develop the desired restored high level (e.g.,2.0 volts) on the bit line and into a selected memory cell. Conversely,an excessively long t₂ timing may not increase the stored high level inthe memory cell as much as it increases the bit line equilibratevoltage, and thus may decrease the high level signal available forsensing, particularly at high VDD.

The timing interval t₃ essentially controls how much charge is sharedbetween the sense amplifier node, the near end and far end of ahigh-going bit line (which typically is moderately resistive), and thememory cell. The resistance of the memory cell access transistor is muchhigher when restoring a high level (due to its lower gate-to-sourcevoltage) than when restoring a low level. The t₃ timing is constrainedby the time needed to write a high voltage into the selected memory cellthrough the array select transistor, the resistive bit line, and furtherthrough the relatively high-resistance memory cell access transistor. Ashort t₃ may result in a worst case memory cell (one located at the“far” end of a bit line, furthest from its bit line sense amplifier)being written to a restored high level which is too low, for a givenamount of “Q” transferred into the sense amplifiers (i.e., for the bitline equilibration voltage which results from the given amount of “Q”).The available signal to be sensed, of course, depends on the highvoltage stored in the memory cell relative to the equilibration voltageof the bit lines and internal bit line sense amplifier nodes.

To appropriately generate these timing intervals t₁, t₂, and t₃ toprovide for highest performance with acceptable operating margins, thelatch timing circuit 930 generates the respective timing signals ST1,ST2, and ST3 using circuitry which includes a word line, a bit linepair, and a sense amplifier, all designed to track (i.e., “mimic”) thedelays of the actual circuitry used in the memory array. Referringspecifically to FIG. 38, some of the major sub-circuits include a“timing” word line TWL, a “timing” bit line pair TBL, TBLB, and a“timing” sense amplifier 966. Other sub-circuits will be introduced anddescribed in the context of describing the operation of the latch timingcircuit 930 through an actual cycle.

Between active cycles, the timing bit line TBL, TBLB is equilibrated bytransistor 936 located at the “near end” of the timing bit line, bytransistor 937 located at the “far end” of the timing bit line, (andoptionally by transistor 965 located at the “mid-point” of the timingbit line), which are each gated by an “early” pulsed-equilibrate signal(a pulsed equilibrate signal that is brought to ground much earlier thannormal pulsed equilibrate signals, such as those for the bit lines andbit line sense amplifiers). The timing sense amplifier 966 is alsoequilibrated between cycles by transistor 947.

The latch formed by inverters 963, 964 is reset between cycles such thatnode 935 is high, which drives nodes 934 and 946 to ground, which“writes” a low level of VSS onto normally-sized memory cell capacitor944 through transistor 943 which is sized like an actual memory cellaccess transistor (i.e., capacitor 944 and transistor 943 replicate thelayers and layout of a normal memory cell), and whose gate terminal isdriven with VPP to approximate the high voltage of a selected word line.Since node 934 is low, transistor 942 is off and the equilibrate levelon the “true” timing bit line TBL is not discharged by transistor 945.Assume that the voltage previously written into normally-sized memorycell capacitor 939 and into normally-sized memory cell capacitor 941,and which is therefore present between cycles, is equal to the restoredhigh level desired for the actual memory array (e.g., about 2.0 volts).All three timing signals ST1, ST2, and ST3 are low between activecycles.

At the end of the previous active cycle, the early pulsed equilibratesignal EP_EQ is brought low after having sufficiently equilibrated thetiming bit line pair TBL, TBLB and timing sense amplifier 966 asdescribed above. This equilibration is completed quickly by usingoversized transistors 936, 937, 947 and, if needed, optional oversizedtransistor 965. A very early timing signal in the clock-to-row timingpath, ECLK, is received by the latch timing circuit 930 to initiate thegeneration of the three timing signals ST1, ST2, and ST3. The particularearly timing signal selected for this role needs to be so early that itis not practical to decode the various control signals to determinewhether the particular active cycle ned not perform the sense andrestore of any sense amplifiers (e.g., such as an idle cycle or a burstcycle) in which case the latch timing circuit 930 need not generate thethree timing signals at all. Rather, the early clock signal ECLK occursevery cycle, and the latch timing circuit 930 goes through its cycle forevery external clock cycle. However, to save power, the three timingsignal outputs ST1, ST2, and ST3 (the earliest of which, ST1, occurssome delay after the early clock signal ECLK), may be enabled ordisabled after determining whether a bit line sense amplifiersense/restore operation is required by the given external cycle.

When an active cycle is initiated, the rising edge of the early clock,ECLK is received by the gate of transistor 961, which grounds node 935(turning off transistor 945), and which “sets” the latch 963, 964 withnode 934 pulled high by inverter 963 (which ensures that nodes 934 and935 are non-overlapping). Consequently, the stored low signal frommemory cell capacitor 944 is then coupled through transistor 943 (sizedto reflect an actual access transistor) and through transistor 942(which is now turned on) and onto the “near end” of the true timing bitline TBL. This stored low, coupled onto the near end of the timing bitline as quickly as possible, is coupled through transistor 934 onto thetrue timing sense amplifier node TSA (also as quickly as possible) tobring the voltage of node TSA down from 1.0 volts to about 0.9 volts(for this exemplary embodiment).

The rising edge of the early clock ECLK is also buffered bylevel-shifting VPP-powered inverter 931 and VPP-powered inverter 932 todrive a “timing” word line TWL, which is driven very early in the activecycle (e.g., about 2 ns earlier than the “selected” word line for thecycle). This timing word line TWL is not necessarily as physically longor as capacitively loaded as an actual word line in the memory array,but is designed (e.g., with the size of inverter 932 scaled downrelative to the actual word line driver) to track the delay of an actualword line. Since the actual memory array word lines are implemented inpolysilicon which is strapped periodically in a metal layer (describedin greater detail below), the timing word line includes a polysiliconportion to track the delays of the word line signal reaching a memorycell located at the “worst case” position (i.e., the memory cell accesstransistor located furthest from the nearest contact to the metalstrap). At the end of the polysilicon tracking portion, the timing wordline TWL is connected to two different memory cells, each identical insize and structure to a normal memory cell in the memory array, whichare both connected to the “far end” of the true timing bit line TBL, andwhich cells, as previously stated, each store a high level.

When the far end of the timing word line TWL exceeds the timing bit lineequilibration voltage by a threshold voltage, the two memory cell accesstransistors 938, 940 begin to conduct, which couples a high level fromeach memory cell capacitor 939, 941 onto the true timing bit line TBL.Since the timing bit line pair TBL, TBLB is also designed to track(i.e., “mimic”) the delays of an actual complementary pair of bit lines,the signal from the “double” (or “2C”) memory cell (i.e., memory cellcapacitors 939, 941) is eventually imparted onto the internal senseamplifier nodes of the timing sense amplifier TSA, TSAB (i.e., “timingsense amplifier,” and “timing sense amplifier bar”), which results inthe true internal node TSA rising from 0.9 volts to 1.1 volts as thehigh level from the “2C” cell overcomes the low level from the “1C” cell(i.e., capacitor 944). This timing bit line arrangement is designed totrack the delay of a signal coupled from a memory cell having a storedhigh level located at the far end of an actual complementary bit linepair, which is the worst case or slowest path.

About the same time as the timing word line is driven high, adifferential amplifier 956 is enabled to amplify the differentialvoltage on the internal timing sense amplifier nodes TSA, TSAB. Whenenabled, this differential voltage is near the maximum negative signaloccurring when the true internal node TSA is about 0.9 volts (from thesignal coupled from memory cell capacitor 944 and before the timing wordline is driven high) and the complement internal node TSAB remain atabout 1.0 volts. When the signal from the “2C” memory cell startsarriving at the sense amplifier 966, the voltage of the true internalnode begins to rise (in approximately an exponential fashion) from 0.9to 1.1 volts, while the complement internal node TSAB remains at about1.0 volts. As the differential signal exceeds zero, the differentialamplifier 956 begins to drive its output high, which is then buffered bya configurable delay circuit 957 to generate the timing signal ST1,which provides an active-high strobe signal to the timing senseamplifier 966 (which signal is coupled through NAND gate 970 andinverter 971 to the gate of N-channel transistor 953, node 974), andwhich signal is inverted by NAND gate 972 to create a complement strobesignal for the timing sense amplifier 966 (which signal is coupled tothe gate of P-channel transistor 948, node 954).

The timing sense amplifier 966 is preferably implemented identically asthe actual bit line sense amplifier circuits (including layoutparasitics), but omits one of the P-channel pull-up transistors (shownby a dotted line 951) and one of the N-channel pull-down transistors(shown by a dotted line 952). This ensures that the timing senseamplifier 966 always latches in the direction to restore a high levelonto the true timing bit line TBL (which then restores the high levelback into the “2C” cell (capacitors 939, 941).

At the same time that the timing sense amplifier 966 begins to latch,the rising edge of timing signal ST1 turns on transistor 962, whichpulls node 934 low (briefly over-powering inverter 963), thereby turningoff transistor 942, all before the timing sense amplifier 966substantially begins to pull up on the true timing bit line TBL. Theinverter 964 then drives node 935 high, which resets the latch 963, 964,and which turns on transistor 945 to restore the low level onto memorycell capacitor 944 (whose other terminal, like all actual memory cellcapacitors, is connected to the “PLATE” voltage). Because nodes 934 and935 are non-overlapping, there can be no current flow from the truetiming bit line TBL through transistor 942 and through transistor 945 toground. Consequently, all of the “Q” provided by the P-channeltransistor 949 (in series with the P-channel latch transistor 948) andconducted onto the true timing bit line TBL results in a higher voltageon the true timing bit line TBL, with no “Q” wasted by conductionthrough transistor 945, nor with any “Q” wasted by conduction throughtransistors 942 and 943 which might otherwise needlessly charge thememory cell capacitor 944 prior to its being written to a low voltage.

The timing sense amplifier 966 is designed to match the delays of anactual bit line sense amplifier in sensing and then, after latching, inrestoring the high and low levels onto the bit lines. When thecomplementary sense enable nodes 974, 954 are driven to cause the timingsense amplifier 966 to begin latching, the true timing bit line TBL isdriven high through transistors 949 and 948, and the complementarytiming bit line TBLB is driven low through transistors 950 and 953.Because the timing bit line is resistive (like the actual bit lines),some time is required to drive the voltage of the far end of the timingbit line to its eventual level. A second differential amplifier 960 isprovided to determine when the voltage of a chosen “tap” along the truebit line TBL (or alternatively, when the true sense amplifier node TSA)exceeds a configurable reference voltage 959 having a nominal value, forexample, equal to 2.0 volts. The timing signal ST2 is then buffered androuted to the memory array to control the turn off time of the PMOSsense enable signals for the selected two rows of regular senseamplifiers within the memory array. It is also inverted (by NOR gate967) and routed to NAND gate 972 to terminate the PMOS sensing in thetiming sense amplifier 966 of the latch timing circuit 930.

The tap position along the true timing bit line TBL is chosen to providethe desired high restore level in the actual bit line sense amplifiers.Such a tap may be easily connected to the timing bit line which isimplemented in a serpentine pattern, as shown, or may be connected tothe true sense amplifier node TSA, as appropriate. The best tapconnection is the one for which the equilibrate voltage establishedfollowing an active cycle varies as little as possible as the operatingsupply voltage VDD varies. The timing interval t₂ should decreasesignicantly as VDD increases. For example, contrast the narrow SEB pulsewidth at a VDD=2.9 volts, as shown in FIG. 25, with the much wider SEBpulse width at a VDD=2.3 volts, as shown in FIG. 26.

The latch timing circuit 930 ensures that the delay between timingsignal ST1 and ST2 (i.e., the PMOS sense timing duration) decreases asthe VDD voltage increases to ensure a written high level which issubstantially independent of VDD, even over process and temperaturecorners. For the exemplary embodiment shown, this is accomplished byusing a “timing” bit line and sense amplifier structure (activatedsubstantially before the main sense amplifiers are activated), anddetecting when the PMOS sensing needs to be turned off to achieve afinal high voltage of about 2.0 volts on the timing sense amplifier andbit line structure. The tap location and configurable reference voltageare set to ensure a written high level on the high bit line (and intothe selected memory cell) of as close to 2.0 volts as possible over aVDD voltage range from 2.3 to 2.9 volts.

The first timing signal, ST1, is also coupled to a second configurabledelay circuit 963, whose output generates, at a configurable time delaylater, the third timing signal, ST3, which is buffered and routed to thememory array to turn off the selected word line and to preferablyterminate the NMOS sense enable. The delay through the configurabledelay circuit 963 may be optimized to be independent of both VDD andtemperature by powering the inverters forming the delay circuit with aregulated supply voltage (coupled to node 964) which is designed toincrease in voltage as temperature increases. The increase in inverterdelay, which otherwise would result from the higher temperature, may bemade to offset the decrease in inverter delay which otherwise wouldresult from the higher “supply voltage” and the resulting inverter delaymay consequently be made constant and independent of both external VDDand temperature. The “configurability” of the delay circuit 963 may beaccomplished, for example, by selectively switching in or out additionalinverter pairs in the path from input to output. Alternatively, theregulated supply voltage coupled to node 964 may be a substantiallyfixed voltage, independent of temperature, to generate a configurabledelay which is independent of VDD, but dependent on temperature. Theconfigurable delay circuit 957, which partially determines the overalldelay from the early clock ECLK to the timing signal ST1, may also bepowered by a regulated supply voltage, as shown, to achieve a constantdelay, independent of temperature and VDD.

The timing sense amplifier 966 is disabled by the timing signal ST2first turning off the PMOS transistor 948, then turning off the NMOStransistor 953. The timing signal ST3 then turns off the timing wordline (by circuitry not shown), thereby trapping the high level on the“2C” memory cells, and the early pulsed equilibrate signal EP_EQ isdriven high. The timing sense amplifier 966 is then equilibrated bytransistor 947, while the timing bit line pair is equilibrated bytransistors 936, 937, and optionally transistor 965 (as describedabove). This EP_EQ signal is then automatically brought low near the endof an active cycle to prepare for a rising edge on the early clocksignal ECLK and the initiation of a new active cycle.

In an alternative embodiment, a suitable PMOS sense enable timing mayalternatively be accomplished using a string of inverters powered at avoltage a fixed amount below VDD, or by other techniques to achieve atiming which is a combination of several variables, such as power supplyvoltage VDD, bandgap voltage, transistor threshold voltage andtransconductance, temperature, or others.

The timing intervals t₁, t₂, and t₃ (respectively created by thecorresponding timing signals ST1, ST2, and ST3) may be collectivelyoptimized on a chip-by-chip basis. In a preferred embodiment, there maybe sixteen different timing settings, each specifying a particularcombination of the t₁, t₂, and t₃ timing intervals, ranging from veryaggressive for highest performance, to very relaxed for highest yield.Referring now to Table 1, the timing setting “1” may provide, forexample, the most aggressive (i.e., shortest) t₁ timing interval, themost aggressive (i.e., shortest) t₂ timing interval, and the mostaggressive (i.e., shortest) t₃ timing interval. The timing setting “16”may provide for the most relaxed t₁ timing interval, the most relaxed t₂timing interval, and the most relaxed t₃ timing interval. Eachincremental timing setting between “1” and “16” is preferably optimizedto incrementally increase, by a similar amount (e.g., by 10 mV), thesignal available at the bit line sense amplifier just before sensing. Toaccomplish this, the timing setting “2” may increase the t, interval by200 ps compared to the “most aggressive” t₁ value of timing setting “1,”while keeping t₂ and t₃ unchanged. A +200 ps increase may be easilyachieved by adding two low-fanout “skewed” inverters to the logic pathsetting the time interval (a skewed inverter being one in which the PMOSpullup transistor and the NMOS pulldown transistor are sized to favordriving its output in a particular direction, rather than sized toprovide similar propagation delay and output rise and fall times whendriving either high or low). A slightly longer increase, such as 220 ps,may be easily achieved by adding two inverters, slightly higher infanout or slightly less skewed (or both), to the logic path setting thetime interval. The timing setting “3” may increase t₃ by 200 ps whilekeeping the same value of the t₁ and t₂ intervals as in timing setting“1.” Each successive low-numbered timing setting increases the value ofone of the three timing intervals t₁, t₂, and t₃ relative to theirvalues in the previous timing setting while keeping the remaining twotiming intervals unchanged. Higher numbered timing settings may increasea given timing interval by increasingly larger amounts to maintain asimilar incremental increase in the signal available at the bit linesense amplifier just before sensing, or may increase more than one ofthe three timing intervals. For example, the timing setting “15” mayincrease t₁ and t₃ each by 400 ps relative to the respective intervalsin timing setting “14” (compared to a 200 ps increase in only the t₃timing interval between timing setting “2” and “3”).

The timing setting “8” is preferably optimized to provide a “nominal”value for each of the three timing intervals t₁, t₂, and t₃ which isexpected to be an appropriate setting for a typical device havingtypical transistor characteristics, typical sense amplifier offsetvoltages, typical bit line resistance, etc., and which is desired toprovide a 100 mV signal in the bit line sense amplifier just beforesensing. Note that these “nominal” values of the timing intervals t₁,t₂, and t₃ are a function of the process corner. Higher bit lineresistance, higher access transistor threshold voltage, or lower VPP,for example, raise the nominal value of the t₁ and t₃ timing intervalswhich are called for by timing setting “8.” For the preferredembodiment, the various timing settings provide a variety of t₁intervals, some shorter than nominal and others longer than nominal, andprovide a variety of t₃ intervals, some shorter and others longer thannominal. But since the duration of the PMOS sensing is so short for thenominal case, for some embodiments the shortest t₂ interval provided isthe “nominal” value, and more relaxed t₂ intervals are provided for inthe timing settings numbered above “8,” as is shown in the Table 1. Forthe embodiment shown in FIG. 38, however, a variety of t₂ intervals,some shorter than nominal and others longer than nominal, may beprovided by altering, for example, the configurable reference voltage959 which is conveyed on wire 962 to the inverting input of thedifferential amplifier 960.

TABLE 1 (for a VDD = 2.3 volts) Desired Signal Timing at Sense SettingInterval t₁ Interval t₂ Interval t₃ Time 1 Fastest Fastest Fastest 30 mV2 +200 ps — — 40 mV 3 — — +200 ps 50 mV 4 +220 ps — — 60 mV 5 — — +220ps 70 mV 6 +240 ps — — 80 mV 7 — — +240 ps 90 mV 8 +270 ps — — 100 mV 9— +200 ps — 110 mV 10 — — +270 ps 120 mV 11 +300 ps — — 130 mV 12 — +240ps — 140 mV 13 — — +300 ps 150 mV 14 +350 ps — — 160 mV 15 +400 ps —+400 ps 170 mV 16 +500 ps — +500 ps 180 mV

During manufacture, this timing setting “8” is configured as the defaultsetting. During a special test mode (for example, at wafer sort) thetiming setting may be temporarily made more or less aggressive todetermine the window of operation for each chip. Some of the memorydevices are found to function correctly with very aggressive timing,while others require more relaxed timing. Then, during the fuse blowingsequence for redundancy, timing fuses may be also blown to permanentlymodify the default strobe timing. The timing setting is preferably setas aggressively as possible to enhance device performance, whilemaintaining adequate sense amplifier signal margins for reliability. Forexample, if a timing setting of “4” is the most aggressive timing forwhich a given device functions without error, then the device may beadvantageously fuse programmed to a timing setting of “6” to ensure someadditional operating margin (the signal to the bit line sense amplifiersincreasing as the timing setting increases). At a later test, such as atfinal test of a packaged device, the test mode may still be entered, andthe timing setting advanced (e.g., by an offset of 2 timing settings)from its then fuse programmed setting (e.g., timing setting “6”) to amore aggressive setting (e.g., resulting in a timing setting of “4”, itspreviously determined most aggressive functional timing setting), inorder to further verify adequate sense amplifier margins on achip-by-chip basis, independent of which actual timing setting was fuseprogrammed into the device.

FIG. 38A is a waveform diagram which illustrates the waveforms for theinternal nodes of the latch timing sense amplifier shown in FIG. 38.When the “low” signal from the “1C” memory cell at the near end of thetiming bit line arrives at the timing sense amplifier, the voltage ofthe true sense amplifier node TSA is brought down by 100 mV (labeled astime 982) relative to the equilibration level of approximately 1.0volts, at which voltage the complement internal timing sense amplifiernode TSAB also remains. Then, when the “high” signal from the “2C”memory cell at the far end of the timing bit line starts to arrive atthe timing sense amplifier, the voltage of the true sense amplifier nodeTSA is brought back high, passing through the voltage of the complementtiming sense amplifier node TSAB (at time 983), and eventually reaches(if its latch enable is delayed long enough) a value of +100 mV relativeto the complement internal node TSAB. Note that when half of the signalfrom the “2C” high memory cell reaches the timing sense amplifier (attime 983), differential amplifier 956 (see FIG. 38) terminates thetiming interval t₁ independent of the particular magnitude of the memorycell capacitance. Referring again to FIG. 38A, when the latch enable forthe timing sense amplifier occurs, the true internal node TSA is drivenhigh (labeled as 980). When it reaches the reference voltage level ofthe configurable reference voltage 959 (labeled at time 985), the timinginterval t₂ is terminated. After the propagation delay through thedifferential amplifier 960 and one or more buffers, the PMOS sensingterminates at a time labeled 986. With the PMOS sensing terminated, thetrue timing sense amplifier node TSA reaches a value (after chargesharing with the bit line) of about 2.0 volts. At the same time, thecomplement internal node TSAB is driven to ground (labeled as 981).Finally, both are equilibrated to about 1.0 volts in preparation for thenext active cycle.

FIG. 39 is a block diagram of such a timing setting control circuitwhich uses an “adder” to generate a signal for selecting one of severalpossible latch timing settings, and which signal may be permanentlymodified by laser fusing to alter the default timing setting, and mayalso be temporarily modified, either before or after laser fusing, byelectrical test signals to alter the timing setting. When in a testmode, the electrical configuration inputs may be provided to the memorydevice by logically enabling all four bytes for any write cycles,independent of the status of the four byte write control inputs, andthen using the four byte write control inputs to specify up to sixteendifferent timing setting offsets, such as −7 through +8.

FIG. 40 is a timing diagram illustrating the general relationshipbetween major timing signals for an array (read or write) operation forvarious embodiments of the memory array described. As is customary insuch “waterfall” charts, the waveforms provide a general representationof the relative timing of the signals shown, and the arrows indicatewhich signal transitions cause corresponding transitions on othersignals (either directly or after some intermediate delay). As all thesignals shown are well described elsewhere herein, this diagram isincluded to provide additional clarity, but need not be discussedseparately.

FIG. 41 is a block diagram of a portion of a memory bank, illustratingthe row strapping, in which alternating metal1 and metal2 word lines areeach strapped to an associated polysilicon word line to reduce word linedelays which would otherwise be present if the word lines wereimplemented only in the polysilicon layer. Such word line straps arepreferably implemented using two different layers of metal (preferablythe two “lowest” layers, metal1 and metal2) in order to match the wordline pitch without requiring any distributed buffers or final decodebuffers Polysilicon layers and interconnections are frequently silicidedto reduce their resistance, and reference herein to “polysilicon” shouldnot be inferred to exclude silicided polysilicon.

The figure is believed to be relatively self-evident, but some featureswhich may not immediately be apparent are worthy of description. Fourword lines WL.0, WL.1, WL.2, and WL.3 are shown. Word lines WL.0 andWL.1 are implemented in the metal1 layer and traverse horizontallyacross the given memory bank (here showing, for example, a small portionof memory bank 561). Word lines WL.2 and WL.3 are implemented in themetal2 layer and also traverse horizontally across the given memorybank. The metal2 word lines are located between the metal1 word lines(on the lower layer of metal below) so that the coupling betweenadjacent word lines is reduced (e.g., word line WL.2 is implemented in ametal2 layer placed largely “between” the underlying metal1 word linesWL.0 and WL.1). As shown in the figure, each of the polysilicon wordlines (which may exist as more than one segments) of a first adjacentpair connect to a respective metal1 word line and each of thepolysilicon word lines of a second (remaining) adjacent pair connect toa respective metal2 word line. In particular, note that the upper pairof polysilicon word lines (1003, 1004) connect to respective word linesWL.0 and WL.1, which are both implemented in metal1. The lower pair ofpolysilicon word lines (1005, 1006) connect to respective word linesWL.2 and WL.3, which are both implemented in metal2. This is done tocoordinate with the row redundancy capability, which replaces word linesin adjacent pairs. Since intra-layer shorts (including poly-to-poly,metal1-to-metal1, and metal2-to-metal2) are more common than inter-layershorts, the efficiency of the limited number of redundant rows isenhanced. The redundant rows replace normal (i.e., non-redundant) rowsin pairs 0,1 or 2,3. With this arrangement, a row-to-adjacent row shortin either metal1, metal2, or poly has a 50% chance of causing a failurein two rows that would be replaced together anyway.

The area labeled 1001 provides a row “strap hole” wherein half of themetal word lines (of a repeating group of four word lines) connect totheir respective polysilicon counterpart. In particular, word line WL.0makes a metal1-to-poly (i.e., polysilicon) contact, and word line WL.2makes a metal2-to-metal1-to-poly (vertical stacked) contact. The othertwo word lines WL.1 and WL.3 traverse through area 1001 withoutcontacting their respective poly word lines. Three of the polysiliconword lines traverse through area 1001 without a break, but thepolysilicon word line WL.1 is discontinuous (i.e., there is a “break” inthe poly) due to the enlarged area of polysilicon required for thecontacts to the poly word line WL.1 and to the poly word line WL.3 toalso fit within this area 1001. However, the two poly segments of WL.1are each contacted by its metal1 counterpart in each adjacent strap holeareas (e.g., area 1002).

Conversely, the area labeled 1002 provides a row “strap hole” whereinthe remaining two metal word lines connect to their respectivepolysilicon counterpart. In particular, word line WL.1 makes ametal1-to-poly contact, and word line WL.3 makes ametal2-to-metal1-to-poly contact. The other two word lines WL.0 and WL.2traverse through area 1002 without contacting their respective poly wordlines. Three of the polysilicon word lines traverse through area 1002without a break, but there is a “break” in the polysilicon word lineWL.2 due to the area required for the two contacts to also fit withinthis area 1002. However, the two poly segments of WL.2 are eachcontacted by its metal2 counterpart in each adjacent strap hole areas(e.g., area 1001).

These two areas 1001, 1002 alternately repeat across each memory bankusing the same interval as the global I/O lines, six of which areindicated in the figure. The read amplifiers used to sense a localoutput line and subsequently drive a global output line may beadvantageously located above the word line strap holes where a break inthe memory cell stepping already occurs to accommodate the row (i.e.,word line) straps. This allows the read amplifier block (e.g., 202, 204of FIG. 4) to more readily be laid out in the center of a group of bitline sense amplifier and column select circuits. As such, the bit linesense amplifier pitch may be slightly less than twice the column pitch(recalling that half of the bit line sense amplifiers are above thearray block and the remaining half below the array block). With thisarrangement, the word line straps are thus located largely beneath thevertically arranged global input and output lines generally traversingoverhead. With the arrangement shown, no memory cell is located morethan the width of 32 columns (bit line pairs) away from its nearestmetal row strap, independent of whether the word line contains aperiodic polysilicon “gap” (such as word line WL.1), or whether the wordline contains no such periodic polysilicon “gaps” (such as word lineWL.0). Of course, with this layout other contacts are needed at the leftand right ends of the memory bank. For example, without contact 1007,the segment 1008 of poly word line WL.2 would be floating. Withoutcontact 1009, the end 1003 of poly word line WL.0 would be 48 columnsfrom its nearest metal row strap, having more than twice the otherwiseworst case distributed RC delay (both R and C increase by a factor of1.5 over the otherwise worst case memory cell being 32 columns away fromits nearest metal row strap).

FIG. 42 is a layout diagram of a portion of a memory bank, illustratingthe row strapping gaps depicted in FIG. 41, in which alternating metal1and metal2 word lines are each strapped to an associated polysiliconword line. The diagram shows areas 1001, 1002 from FIG. 41 (and a smallportion of the surrounding layout), although a number of columns (bitline pairs) have been removed to fit the size of the page without lossof comprehension. In the FIG. 42, a heavy-lined (“dark”) square contact,such as contact 1020, is a metal2-to-metal1-to-poly stacked verticalcontact, whereas a lightly-lined square contact, such as contact 1021,is a metal-to-poly contact. As the structure of this layout wasrelatively well represented in FIG. 41, reference should be made to theearlier description to assist in understanding. As a further point ofclarity, a bit line cross-over structure, labeled 1022, is shown whichuses metal1 and metal2 to cross a polysilicon bit line pair over eachother and over another polysilicon bit line pair therebetween. Alsoshown are two dummy rows 1023, 1024 of the guard cells 801 previouslyshown in FIG. 36. A pair of polysilicon dummy bit lines is alsoimplemented on either side of each row strap hole to providephotolithographic guard cells at the side of each arrayed group ofmemory cells (e.g., dummy bit line pair 1025, 1026 on the left of area1002, and dummy bit line pair 1027, 1028 on the right of area 1002). Forclarity, the many layers not associated with the word line straps or thebit line cross-over structures are not included in the figure.

FIG. 43 is a schematic diagram of another embodiment of a column decodearrangement for coupling a selected sense amplifier through a pair oflocal I/O lines to a pair of global output lines when reading, and forcoupling a pair of global input lines through the pair of local I/Olines to the selected sense amplifier when writing, with the even columnaddresses selecting a sense amplifier below the array block, and the oddcolumn addresses selecting a sense amplifier above the array block, bothof which are coupled to the same set of global input/global outputlines. Such an embodiment (using 1 set of GOUT lines to serve both thesense amplifiers above the array block and below the array block) may beadvantageous if no burst mode need be provided, particularly if thelayout is too squeezed to fit two pairs of global input line wires andtwo pairs of global output line wires between the massive vertical VDDand VSS power supply wires.

A preferred embodiment of the memory device receives an external clocksignal EXT_CLK and a variety of other control signals, including aread/write control R/W#, an advance/load control ADV/LOAD #, a chipenable CE, and a clock enable CLKEN. Table 2 describes the internaloperation performed in response to various combinations of theseexternal control signals.

TABLE 2 (assumes CLKEN is valid) CE# ADV/LOAD# R/W# Action Taken Valid(L) LOAD (L) R (H) Start Read (load cycle) Valid (L) LOAD (L) W (L)Start Write (load cycle) (Don't care) ADV (H) (Don't Care) Continue (R,W, or Stop) Invalid (H) LOAD (L) (Don't care) Stop

In the preferred embodiment, the memory device includes refresh controlcircuitry for automatically performing internal refresh of the memoryarray without user intervention. A separate refresh control circuit isprovided for each memory bank, which are each configured to request arefresh request, for its respective memory bank, every 256 externalclock cycles. When a refresh request for a memory bank is generated byits refresh control circuitry, it is immediately performed if thatmemory bank is not otherwise occupied with an internal “load” cycle(i.e., a load read or write cycle to an external address, or anautomatic load read or write cycle to an internally generated address tocontinue a burst read or write). If the memory bank happens to be busywith such a load cycle and is unable to perform the refresh whenrequested, the request is queued and performed the first cycle that thememory bank is available (i.e., not executing an internal load cycle).

Unlike earlier devices, such a “hidden” refresh cycle can fully completein just one cycle, and another internal load cycle can begin in the samememory bank on the very next external cycle, if required, in accordancewith the next command and address received. Thus, even if the memorybank is free for only one cycle, a hidden refresh operation can be fullyperformed and the memory bank ready on the very next cycle to accept anexternal load cycle with full confidence. This makes such hidden refreshcycles possible totally under internal control, with no userinteraction, because there is no risk that an external cycle will bereceived that cannot be carried out because the memory bank is stillbusy performing an internal refresh cycle. For the same reason, there isno need for a “busy” signal to alert the user when an internal refreshcycle is in progress.

The respective counters within each refresh control circuit are offsetso that every 64 external cycles one of the four memory banks adds “1”to its number of queued or pending refresh requests. Whenever the numberin the queue exceeds zero, that memory bank attempts a refresh on everycycle. Therefore, on the first available cycle after the queue isincremented, the request is retired (i.e., the refresh cycle isperformed) and the count in the queue is decremented. No further refreshrequests are pending in the queue when it decrements to its normal stateof zero. For every 256 external clock cycles, as long as a memory bankis free for just one cycle, a hidden refresh can be performed withouthaving the queued number of refresh requests increase. In the memorydevice embodiment described, two particular address bits (e.g., the A2and A1 bit) choose which memory bank is addressed (recall that the LSB,address bit A0, chooses the between the upper or lower 36-bit wordaccessed from a single memory bank). If the user arranges thesignificance of the external address bits to use the same significanceas the memory device, then for any reasonable addressing sequence whichaccesses small or large blocks of memory, or even those which randomlyaddress various addresses, the lower significance address bits arealmost assuredly changing frequently. Consequently, as the memory deviceresponds to the particular sequence of addresses, all four memory banksare likely addressed with some reasonable frequency (thus ensuring thateach memory bank is free reasonably often, such as, on average, 192 outof 256 cycles if all cycles are load read or load write cycles, and evenmore if some cycles are idle or burst cycles). The likelihood of asingle memory bank being busy for 256 straight cycles is very low.Moreover, the likelihood of a single memory bank being busy foradditional multiples of 256 straight cycles is extremely low.

A refresh request does not need to be retired before the next refreshrequest arrives. Up to 64 refresh requests may be placed into therefresh queue. A refresh flag is provided in the extremely unlikelyevent that the internal refresh for a particular memory bank gets veryfar behind its desired rate (i.e., if too many refresh requests arequeued). If the refresh queue is nearly full, the refresh flag isasserted. For example, if a particular memory bank has not performed arefresh cycle in 61×256 external clock cycles, the refresh flag isasserted. However, even if a memory bank gets seriously “behind” and hasqueued up many refresh requests, if the memory bank thereafter is freefor at least two cycles out of every 256 external clock cycles, refreshrequests will be retired from the queue faster than new ones will beadded, and the device will eventually “catch up.” It is for this reason,along with the depth of the refresh queue, that the refresh flag isseldom, if ever, expected to be asserted (provided the user assigns thesignificance of the address bits as described).

The refresh counters and control circuits are also arranged to ensurethat no more than three memory banks can perform an internal arrayoperation at the same time (one performing an internal load cycle, andtwo each performing a hidden refresh cycle) to reduce the worst casecurrent transients (i.e., power supply noise) that would occur if allfour memory banks were simultaneously active.

The choice of counting clock cycles to determine when to refresh, ratherthan elapsed time, has several advantages. First, it is easy to do. Theclock exists and is always running (as required by the phase-lockeddelay line previously described). Second, the refresh interval requiredfor proper operation is more or less proportional to the operating cycletime. As the operating frequency increases (i.e., cycle time decreases),more minority carriers are injected into the substrate, increasing theleakage of the memory cells, thus decreasing the data retention time ofthe memory cells. Consequently, faster cycling (i.e., shorter cycletimes) requires more frequent refreshing. Furthermore, anothersignificant source of cell leakage is the sub-threshold conduction ofthe access transistors on unselected rows of memory cells. Thissub-threshold conduction only occurs when the bit line (or complementbit line), to which the unselected memory cell is attached, is biased atVSS, which only occurs for about 2 ns per cycle. This component ofmemory cell leakage is directly proportional to frequency. Again, higherfrequency operation requires more frequent refreshing.

In the preferred embodiment, the memory device supports burst mode forboth read and write cycles. For example, a burst mode read sequence toread four consecutively-addressed 36-bit single words is accomplished bypresenting the address of the first of the four words, and initiating afirst “load” cycle as indicated in Table 2 (which, as describedpreviously, drives the corresponding output data for the first wordduring a subsequent external cycle). The next three cycles are theninitiated as “continue” or “advance counter” cycles, and no addressesneed be preseted to the memory device. Rather, the memory deviceincrements the address received from the load cycle (in either of twowell-known counting orders) to provide the proper addresses for the nextthree cycles. Consequently, the external memory bus is available forother use, and is not required to service the memory device inperforming the three successive burst mode cycles.

The above description of burst mode is cast from the perspective of theuser of the memory device, and is not particularly instructive of theinternal operation of the memory device. For example, such a burst modecapability may be (and usually is) supported by a memory device whichmaintains an internal address counter (which is initialized by theaddress of the load cycle, and incremented for each successive burstmode cycle), and which performs a full 36-bit access into the memoryarray for each of the four burst mode cycles (which, in effect, is anexternal load cycle for the first cycle, and is an “internal load cycle”for each of the following three burst mode cycles). However, in thepreferred embodiment, the internal data path to and from the memoryarray is a 72-bit wide path corresponding to two 36-bit single wordswhose addresses differ only in the LSB. Taking advantage of thiscapability, upon receipt of the external load cycle (which starts theburst), the memory device may perform an “internal” load cycle whichretrieves the full 72-bit double word which includes the 36-bit singleword addressed by the external load cycle, as well as the “other” 36-bitword whose address differs only in that its least significant addressbit is opposite that of the addressed word. The addressed 36-bit word isdirected to the output buffers as normal, and the second 36-bit word isstored internally in registers located outside the array. Consequently,in the next cycle, the stored, second 36-bit word may be retrieved andprovided to the output buffers (assuming the burst counting sequencecorresponds to the address of the stored second word) and the memoryarray need not perform another load cycle to retrieve the data. Insteadthe memory array can remain inactive, thus saving considerable power, ormay be called upon to perform an internal refresh cycle (i.e., a“hidden” refresh cycle) at the same time that the memory array “appears”to be occupied with supporting the burst mode cycle. When operating inburst mode, the already remote chance that a given memory bank is unableto keep up with the internally generated refresh requests all butdisappears.

If all, or at least many, memory cycles are burst mode cycles, theninternally accessing the full 72-bit double word and providing 36 of thebits to the output buffers during a first cycle, and providing theremaining 36 bits to the output buffers during the next cycle (assumingit is a burst cycle), results in far less power consumption than if twoseparate memory array operations were performed, each accessing only a36-bit single word and providing it to the output buffers. However, ifthe address of each successive external memory cycle is unpredictable,rather than sequential in nature, and seldom addresses through at leasta group of two consecutive addresses, then the “other” 36-bit word whichis accessed during the 72-bit internal load cycle and stored inanticipation of the next cycle being a burst cycle, is frequently (oralways) unused because the next address does not correspond to thisstored word. In such a case, the memory device would be moreadvantageously configured to only access 36-bit words during an internalload cycle, thus saving the power consumption otherwise required to readand save the second word (which is ignored when the next external readcycle addressing a different word is received).

To provide for both modes of operation, in the preferred embodiment thememory device powers up in the non-burst mode. A burst mode flip-flop isreset upon power-up, and all internal load cycles are consequently36-bit cycles. The first time a user initiates an ADVANCE cycle after aLOAD READ cycle (in other words, the first time the user tries toutilize a burst mode read cycle), the memory device computes theincremented address accordingly, and internally performs a second 36-bitinternal load cycle to access the other half of the double word from thememory array. The memory device then correctly provides the read data tothe output pins at the correct time. However, this first burst modecycle also sets the burst mode flip-flop so that all future internalload read cycles are 72-bit load cycles, and consequently future burstmode read cycles may use the stored “second” 36-bit word retrievedduring the 72-bit load read cycle. Such an internal 72-bit load readcycle uses approximately 20% more power than an internal 36-bit loadread cycle. However, if all cycles are burst of four read cycles, onlyhalf as many internal load cycles need be performed. With half thecycles using 120% the power of a single 36-bit load read cycle, and theother half using zero memory array power, the average power consumed isonly 60% of that used by consecutive 36-bit internal load read cycles(i.e., without the burst mode flip-flop being set).

In a burst mode sequence, the first 36-bit word in the consecutivesequence is retrieved from the memory bank corresponding to the readaddress using a 72-bit internal load cycle. If the burst mode continuesto read a second word of the burst, the corresponding data may havealready been retrieved by the first 72-bit internal load cycle(depending on the counting sequence, and the particular startingaddress), and can be provided to the data output buffers withoutperforming another internal load cycle. Assume for a moment such a case.Then, if the burst mode continues to read a third word of the burst, asecond internal load cycle is necessary and is automatically initiatedto retrieve the next 72 bits of data, all transparent to the user. Theaddress decoding may be arranged so that these next two 36-bit words ofdata in the consecutive sequence may correspond to another column oranother word line in the same memory bank as the first two 36-bit words,or may be arranged to correspond to the same row and column addresswithin another memory bank (e.g., an adjacent memory bank).

Conversely, the corresponding data may not have already been retrievedby the first 72-bit internal load cycle (depending on the countingsequence, and the particular starting address) when the burst modecontinues to read the second word of the burst. In such a case, a secondinternal load cycle is necessary to retrieve even the second 36-bitword, and is automatically initiated (an “auto-load during a burstcycle”) to retrieve the next 72 bits of data, all transparently to theuser. Then, if the burst mode read continues to read a third word of theburst, the corresponding data has already been retrieved by the second72-bit internal load cycle, and can be provided to the data outputbuffers without performing another internal load cycle. With thissequence, the fourth and final word of the burst was obtained as theunused half of the original 72-bit load read cycle, and no additionalmemory array cycle is required to output that data. Thus, an internalload cycle is automatically performed, without user interaction,whenever the burst mode counting sequence increments to an address forwhich the corresponding data has not already been retrieved by a 72-bitinternal load cycle.

Since any internal load cycle, including an automatically initiatedinternal load cycle, as well as any burst mode ADVANCE cycle can fullycomplete in just one external clock cycle, the burst mode sequence isfully interruptible after any arbitrary cycle of the burst, and a newinternal load read or load write cycle, having a totally arbitraryaddress, can be executed during the very next external clock cycle.

The burst mode thus far has been described largely in the context ofread cycles. The write cycle merging capability already describedprovides the internal capabilities in the data path which are necessaryto support burst mode write cycles. In particular, recall that twoconsecutive write cycles, writing data to two 36-bit words whoseaddresses differ only in the LSB, are merged (whether or not they werereceived in a burst mode) so that only one internal 72-bit writeoperation is performed. The address incrementing capability, alreadydescribed above for read burst cycles, is also required to support burstmode write cycles because, like burst mode read cycles, the user neednot present the write address on subsequent burst mode cycles after thefirst “LOAD WRITE” cycle.

In the preferred embodiment a burn-in mode is provided which dispenseswith most of the internal timing, and drives every other word line inevery array block (in all four memory banks) to a DC high voltage (whichis adjustable, and usually set to a higher voltage than the normal VPPvoltage), enables the column decoders in each hole (including both theleft and right decoded write signals), latches the bit line senseamplifiers in every hole, and holds all these signals for an entiresecond. As a result, every word line in the entire memory device is at avoltage opposite that of both of its neighboring word lines, every bitline is driven to a voltage opposite that of both respective neighboringbit lines for at least half the time with the proper choice of datapattern, and one-fourth of the memory cells have a voltage stress placedacross its dielectric, for substantially the entire duration of eachone-second long cycle. This affords a significant decrease in theburn-in times required to adequately stress the device for at least tworeasons. First, the cycle time achievable during burn-in is frequentlymuch longer than during normal operation. When testing a self resettingdevice at such slow cycle times, the device only infrequently performsan active cycle, and otherwise remains in a reset or precharge statemost of the time. By holding the signals stated above for the entirevery long active cycle, the desired voltage stress is applied forsubstantially the entire active cycle. This increases the duty cycle ofactive cycles from about 2 ns per 1000 ns for a typical 1 MHz clockduring burn-in, to almost 100% duty cycle, which is a factor of 500improvement! Second, this technique described above allows so much moreof the circuit elements to be stressed at the same time than normallyachievable if the decode functions were to be operating normally. Thisincreases the number of word lines being stressed simultaneously from 1(i.e., one word line in one memory bank) to 8192 word lines (i.e., halfof the 4096 word lines in each of the four memory banks), a factor of8000 improvement. The stress time is thus increased by a factor of(500)(8192)=16,385,000. Every 2 seconds of burn-in using this modeaccomplishes about the same stress to the memory array as 1 year ofburn-in under normal operation at the very low operating frequency of 1MHz.

Using the teachings described above, the exemplary dynamic memory arrayarchitecture described above affords random access cycles (eachrequiring a new row access) at a sustained rate in excess of 200 MHzoperation for memory devices tolerating aggressive t₁, t₂, and t₃timing, even when each new row access is within the same array block ofthe same memory bank!

The many aspects, features, and advantages of the present invention areconveyed herein by describing several exemplary embodiments of ahigh-performance DRAM memory device. In some instances, simplified blockdiagrams and schematics are shown, particularly when the key concepts,features, or implementation details may be more easily communicated. Inother cases, more complete schematics are shown if helpful to impart amore complete understanding of the invention or to better appreciate itsnuances. One skilled in the art will recognize the many teachings ofthis disclosure and be able to apply these teachings to additionalembodiments and, indeed, to other kinds of devices, as well, withoutdeparting from the teachings of this disclosure. For example, theteachings of this disclosure may also be advantageously applied tomemory arrays incorporated within an integrated circuit that includes aprocessor, such as an integrated processor (e.g., microprocessor)circuit including an embedded dynamic memory array. Consequently, thescope of the invention should not be inferred as being limited by theexemplary embodiments described herein, but rather should be viewed asteaching in the art far greater than just these exemplary embodiments.Accordingly, other embodiments, variations, and improvements notdescribed herein are not necessarily excluded from the scope of theinvention.

General Nomenclature and Terminology Usage

Regarding terminology used herein, it will be appreciated by one skilledin the art that any of several expressions may be equally well used whendescribing the operation of a circuit including the various signals andnodes within the circuit. Any kind of signal, whether a logic signal ora more general analog signal, takes the physical form of a voltage level(or for some circuit technologies, a current level) of a node within thecircuit. It may be correct to think of signals being conveyed on wiresor buses. For example, one might describe a particular circuit operationas “the output of circuit 10 drives the voltage of node 11 toward VDD,thus asserting the signal OUT conveyed on node 11.” This is an accurate,albeit somewhat cumbersome expression. Consequently, it is well known inthe art to equally describe such a circuit, operation as “circuit 10drives node 11 high,” as well as “node 11 is brought high by circuit10,” “circuit 10 pulls the OUT signal high” and “circuit 10 drives OUThigh.” Such shorthand phrases for describing circuit operation are moreefficient to communicate details of circuit operation, particularlybecause the schematic diagrams in the figures clearly associate varioussignal names with the corresponding circuit blocks and node names. Forconvenience, an otherwise unnamed node conveying the CLK signal may bereferred to as the CLK node. Similarly, phrases such as “pull high,”“drive high,” and “charge” are generally synonymous unless otherwisedistinguished, as are the phrases “pull low,” “drive low,” and“discharge.” It is believed that use of these more concise descriptiveexpressions enhances clarity and teaching of the disclosure. It is to beappreciated by those skilled in the art that each of these and othersimilar phrases may be interchangeably used to describe common circuitoperation, and no subtle inferences should be read into varied usagewithin this description.

As an additional example, a logic signal has an active level and aninactive level (at least for traditional binary logic signals) and theactive and inactive levels are sometimes also respectively called activeand inactive “states.” The active level for some logic signals is a highlevel (i.e., an “active-high” signal) and for others is a low level(i.e., an “active-low” signal). A logic signal is “asserted” or“activated” when driven to the active level. Conversely, a logic signalis “de-asserted” or “de-activated” when driven to the inactive level. Ahigh logic level is frequently referred to as a logic “1” and a lowlogic level is frequently referred to as a logic “0” (at least forpositive logic).

Frequently logic signals are named in a fashion to convey which level isthe active level. For example, CLKEN is commonly used to name anactive-high clock enable signal, because the true polarity is implied inthe name. Conversely, CLKENB, /CLKEN, CLKEN#, CLKEN*, CLKEN_L, CLKEN_C,or #CLKEN are commonly used to name an active-low clock enable signal,because one of the many common expressions indicating the complementpolarity is used in the name. Complementary pairs of signals or nodenames, such as true and complement clock lines, and true and complementbit lines within a column of a memory array, are frequently named toclarify the polarity of both nodes or signals (e.g., BL3T and BL3C;BL6_T and BL6_C) and in other instances, only the complement polaritymay be indicated in the names (e.g., CLK and CLK#, or BL and BLB). Instill other cases, two “complementary” signals are both inactive at onestate (e.g., inactive low), and only one is driven to an active level toconvey the polarity of the signal. For example, two complementaryaddress lines (e.g., A2T and A2C) are both inactive low during aquiescent portion of a cycle. Later, A2T is driven high to indicate thatthe received address A2 is high (A2=H). Alternatively, A2C is drivenhigh to indicate that the address received is low (A2=L). It is to beappreciated by those skilled in the art that these and other similarphrases may be used to name the signals and nodes. The schematicdiagrams and accompanying description of the signals and nodes should incontext be clear.

A transistor may be conceptualized as having a control terminal whichcontrols the flow of current between a first current handling terminal(or current carrying terminal) and a second current handling terminal.An appropriate condition on the control terminal causes a current toflow from/to the first current handling terminal and to/from the secondcurrent handling terminal (for typical operating voltages of the firstand second current handling terminals). In a bipolar NPN transistor, thefirst current handling terminal may be deemed the emitter, the controlterminal deemed the base, and the second current handling terminaldeemed the collector. A sufficient base current into the base causes acollector-to-emitter current to flow (for typical collector-to-emitteroperating voltages). In a bipolar PNP transistor, the first currenthandling terminal may be deemed the emitter, the control terminal deemedthe base, and the second current handling terminal deemed the collector.A sufficient base current exiting the base causes anemitter-to-collector current to flow (for typical collector-to-emitteroperating voltages).

An MOS transistor may likewise be conceptualized as having a controlterminal which controls the flow of current between a first currenthandling terminal and a second current handling terminal. Although MOStransistors are frequently discussed as having a drain, a gate, and asource, in most such devices the drain is interchangeable with thesource. This is because the layout and semiconductor processing of thetransistor is symmetrical (which is typically not the case for bipolartransistors). For an N-channel MOS transistor, the current handlingterminal normally residing at the higher voltage is customarily calledthe drain. The current handling terminal normally residing at the lowervoltage is customarily called the source. A sufficiently high voltage onthe gate (relative to the source voltage) causess a current to thereforeflow from the drain to the source (provided the respective voltage ofthe drain and source are different). For an enhancement mode N-channeldevice, a positive gate-to-source voltage greater than the thresholdvoltage (including body effect) is sufficient. The source voltagereferred to in N-channel MOS device equations merely refers to whichevercurrent handling terminal has the lower voltage at any given point intime. For example, the “source” of the N-channel device of abi-directional CMOS transfer gate depends on which side of the transfergate is at the lower voltage. To reflect this symmetry of most N-channelMOS transistors, the control terminal may be deemed the gate, the firstcurrent handling terminal may be termed the “drain/source”, and thesecond current handling terminal may be termed the “source/drain”. Sucha description is equally valid for a P-channel MOS transistor, since thepolarity between drain and source voltages, and the direction of currentflow between drain and source, is not implied by such terminology.Alternatively, one current handling terminal may arbitrarily deemed the“drain” and the other deemed the “source”, with an implicitunderstanding that the two are not distinct, but interchangeable.

Regarding power supplies, a single positive power supply voltage (e.g.,a 2.5 volt power supply) used to power a circuit is frequently named the“VDD” power supply. In an integrated circuit, transistors and othercircuit elements are actually connected to a VDD terminal or a VDD node,which is then operably connected to the VDD power supply. The colloquialuse of phrases such as “tied to VDD” or “connected to VDD” is understoodto mean “connected to the VDD node”, which is typically then operablyconnected to actually receive the VDD power supply voltage during use ofthe integrated circuit.

The reference voltage for such a single power supply circuit isfrequently called “VSS.” Transistors and other circuit elements areactually connected to a VSS terminal or a VSS node, which is thenoperably connected to the VSS power supply during use of the integratedcircuit. Frequently the VSS terminal is connected to a ground referencepotential, or just “ground.” Describing a node which is “grounded”by aparticular transistor (unless otherwise defined) means the same as being“pulled low” or “pulled to ground” by the transistor.

Generalizing somewhat, the first power supply terminal is frequentlynamed “VDD”, and the second power supply terminal is frequently named“VSS.” Both terms may appear either using subscripts (e.g., VDD) or not.Historically the nomenclature “V_(DD)” implied a DC voltage connected tothe drain terminal of an MOS transistor and V_(SS) implied a DC voltageconnected to the source terminal of an MOS transistor. For example, oldPMOS circuits used a negative VDD power supply, while old NMOS circuitsused a positive VDD power supply. Common usage, however, frequentlyignores this legacy and uses VDD for the more positive supply voltageand VSS for the more negative (or ground) supply voltage unless, ofcourse, defined otherwise. Describing a circuit as functioning with a“VDD supply” and “ground” does not necessarily mean the circuit cannotfunction using other power supply potentials. Other common power supplyterminal names are “VCC” (a historical term from bipolar circuits andfrequently synonymous with a +5 volt power supply voltage, even whenused with MOS transistors which lack collector terminals) and “GND” orjust “ground.”

What is claimed is:
 1. In an integrated circuit including a differentialtransistor pair having a common node connecting a respective firstcurrent handling terminal of each transistor within the pair, a methodof providing a tail current for the differential transistor paircomprising: precharging a capacitive device having a first terminal anda second terminal; providing a current path from the common node of thedifferential transistor pair to the first terminal of the capacitivedevice; and driving, with a substantially constant first current, thevoltage of the second terminal of the precharged capacitive device in adirection to capacitively couple the first terminal and, by way of thecurrent path, the common node to respective voltages sufficient inmagnitude and polarity to cause a current, substantially equal to thefirst current, to flow collectively through one or both transistors ofthe differential transistor pair; wherein the first current flows as adisplacement current through the capacitive device and is projected ontothe common node, thereby providing a dynamic tail current for thedifferential transistor pair substantially equal in magnitude to thefirst current.
 2. A method as in claim 1 wherein: the voltage of thesecond terminal of the capacitive device, while the first current flowsthrough the capacitive device as a displacement current, changes at arate that substantially exceeds the rate of voltage change of the firstterminal of the capacitive device.
 3. A method as in claim 1 wherein:each transistor of the differential transistor pair comprises a fieldeffect transistor; and the common node comprises a common-source node.4. A method as in claim 1 wherein: each transistor of the differentialtransistor pair comprises a bipolar junction transistor; and the commonnode comprises a common-emitter node.
 5. A method as in claim 1 wherein:while the first current flows through the capacitive device as adisplacement current, the first terminal of the capacitive deviceassumes a voltage at which the current collectively flowing through oneor both transistors of the differential transistor pair remainssubstantially equal to the first current.
 6. A method as in claim 5wherein: for certain values of input voltages applied to thedifferential transistor pair, certain transistor characteristics of thedifferential transistor pair, and certain magnitudes of the firstcurrent, the voltage assumed by the first terminal of the capacitivedevice is outside a range of voltages bounded by the highest powersupply voltage and the lowest power supply voltage operably received bythe integrated circuit.
 7. A method as in claim 5 wherein: for certainvalues of input voltages applied to the differential transistor pair,certain transistor characteristics of the differential transistor pair,and certain magnitudes of the first current, the voltage assumed by thefirst terminal of the capacitive device is within a range of voltagesbounded by the highest power supply voltage and the lowest power supplyvoltage operably received by the integrated circuit.
 8. A method as inclaim 1 wherein: the capacitive device is disposed in close localproximity to, and uniquely associated with, the differential transistorpair.
 9. A method as in claim 1 wherein: the capacitive device isdisposed in close local proximity to, and also uniquely associated with,said differential transistor pair and another differential transistorpair; and each differential transistor pair includes a respective gatingcircuit connecting the respective common node thereof to the firstterminal of the capacitive device.
 10. In an integrated circuitincluding a differential transistor pair having a common node connectinga respective first current handling terminal of each transistor withinthe pair, a method of providing a tail current for the differentialtransistor pair comprising: precharging a capacitive device having afirst terminal and a second terminal; providing a current path from thecommon node of the differential transistor pair to the first terminal ofthe capacitive device; and driving, with a first current, the voltage ofthe second terminal of the precharged capacitive device in a directionto capacitively couple the first terminal and, by way of the currentpath, the common node to respective voltages sufficient in magnitude andpolarity to cause a current, substantially equal to the first current,to flow collectively through one or both transistors of the differentialtransistor pair; wherein the first current flows as a displacementcurrent through the capacitive device and is projected onto the commonnode, thereby providing a dynamic tail current for the differentialtransistor pair substantially equal in magnitude to the first current;and wherein the current path consists of a direct wired connectionbetween the common node and the first terminal of the capacitive device.11. A method as in claim 1 wherein: the current path includes at leastone transistor connecting the common node to the first terminal of thecapacitive device, which at least one transistor is enabled to conductcurrent therebetween at least while driving the second terminal of thecapacitive device.
 12. A method as in claim 11 wherein: while drivingthe second terminal of the capacitive device, the current path includesno device configured to function as a constant current source; and themagnitude of the displacement current is controlled by a circuit drivingthe second terminal of the capacitive device with the first current. 13.A method as in claim 11 wherein: while driving the second terminal ofthe capacitive device, the current path includes at least one deviceconfigured to function as a constant current source; and the magnitudeof the displacement current is controlled by the constant currentsource.
 14. A method as in claim 1 further comprising: developing adifferential output signal at respective second current handlingterminals of the differential transistor pair while the displacementcurrent through the capacitive device provides a tail current for thedifferential transistor pair; and utilizing the differential outputsignal in a subsequent circuit coupled to the differential transistorpair.
 15. A method as in claim 14 wherein the differential transistorpair is incorporated within a linear amplifier circuit.
 16. A method asin claim 15 wherein the integrated circuit further comprises: a pair ofload devices, each respectively coupled between the respective secondcurrent handling terminal of each transistor within the differentialtransistor pair and a power supply terminal.
 17. In an integratedcircuit including a differential transistor pair having a common nodeconnecting a respective first current handling terminal of eachtransistor within the pair, a method of providing a tail current for thedifferential transistor pair comprising: precharging a capacitive devicehaving a first terminal and a second terminal; providing a current pathfrom the common node of the differential transistor pair to the firstterminal of the capacitive device; and driving, with a controlled firstcurrent, the voltage of the second terminal of the precharged capacitivedevice in a direction to capacitively couple the first terminal and, byway of the current path, the common node to respective voltagessufficient in magnitude and polarity to cause a current, substantiallyequal to the first current, to flow collectively through one or bothtransistors of the differential transistor pair; wherein the firstcurrent flows as a displacement current through the capacitive deviceand is projected onto the common node, thereby providing a dynamic tailcurrent for the differential transistor pair substantially equal inmagnitude to the first current; developing a differential output signalat respective second current handling terminals of the differentialtransistor pair while the displacement current through the capacitivedevice provides a tail current for the differential transistor pair; andutilizing the differential output signal in a subsequent circuit coupledto the differential transistor pair; and wherein the differentialtransistor pair is incorporated within a latching amplifier circuithaving a separate latch enable input.
 18. In an integrated circuitincluding a differential transistor pair having a common node connectinga respective first current handling terminal of each transistor withinthe pair, a method of providing a tail current for the differentialtransistor pair comprising: precharging a capacitive device having afirst terminal and a second terminal; providing a current path from thecommon node of the differential transistor pair to the first terminal ofthe capacitive device; and driving, with a controlled first current, thevoltage of the second terminal of the precharged capacitive device in adirection to capacitively couple the first terminal and, by way of thecurrent path, the common node to respective voltages sufficient inmagnitude and polarity to cause a current, substantially equal to thefirst current, to flow collectively through one or both transistors ofthe differential transistor pair; wherein the first current flows as adisplacement current through the capacitive device and is projected ontothe common node, thereby providing a dynamic tail current for thedifferential transistor pair substantially equal in magnitude to thefirst current; developing a differential output signal at respectivesecond current handling terminals of the differential transistor pairwhile the displacement current through the capacitive device provides atail current for the differential transistor pair; and utilizing thedifferential output signal in a subsequent circuit coupled to thedifferential transistor pair; wherein the differential transistor pairis incorporated within a latching amplifier circuit having no separatelatch enable input; and wherein the latching amplifier circuit latchesas a result of the differential output signal developed by thedifferential transistor pair.
 19. A method as in claim 18 wherein theintegrated circuit further comprises: a pair of cross-coupled invertersconfigured as a latch, each inverter's output respectively connected tothe other inverter's input and also connected to a respective secondcurrent handling terminal of the transistors within the differentialtransistor pair; wherein the displacement current flowing through thecapacitive device and providing a tail current for the differentialtransistor pair provides a strobe signal for the latch.
 20. In anintegrated circuit including a differential transistor pair having acommon node connecting a respective first current handling terminal ofeach transistor within the pair, a method of providing a tail currentfor the differential transistor pair comprising: precharging acapacitive device having a first terminal and a second terminal;providing a current path from the common node of the differentialtransistor pair to the first terminal of the capacitive device; anddriving, with a controlled first current, the voltage of the secondterminal of the precharged capacitive device in a direction tocapacitively couple the first terminal and, by way of the current path,the common node to respective voltages sufficient in magnitude andpolarity to cause a current, substantially equal to the first current,to flow collectively through one or both transistors of the differentialtransistor pair; wherein the first current flows as a displacementcurrent through the capacitive device and is projected onto the commonnode, thereby providing a dynamic tail current for the differentialtransistor pair substantially equal in magnitude to the first current;developing a differential output signal at respective second currenthandling terminals of the differential transistor pair while thedisplacement current through the capacitive device provides a tailcurrent for the differential transistor pair; and utilizing thedifferential output signal in a subsequent circuit coupled to thedifferential transistor pair; and wherein the differential transistorpair is incorporated within an external input buffer of the integratedcircuit.
 21. A method as in claim 20 wherein: the integrated circuitincludes a memory array.
 22. A method as in claim 20 wherein: thedriving step is controlled to occur at a time relative to an externalclock signal received by the integrated circuit.
 23. A method as inclaim 22 wherein: the external input buffers of the integrated circuitare strobed at a time relative to an external clock signal received bythe integrated circuit.
 24. A method as in claim 20 further comprising:coupling an external input signal to a control terminal of onetransistor of the differential transistor pair; and coupling a referencevoltage to a control terminal of the other transistor of thedifferential transistor pair.
 25. A method as in claim 20 furthercomprising: coupling an external input signal to a control terminal ofone transistor of the differential transistor pair; and coupling acomplementary external input signal to a control terminal of the othertransistor of the differential transistor pair.
 26. In an integratedcircuit including a differential transistor pair having a common nodeconnecting a respective first current handling terminal of eachtransistor within the pair, a method of providing a tail current for thedifferential transistor pair comprising: precharging a capacitive devicehaving a first terminal and a second terminal; providing a current pathfrom the common node of the differential transistor pair to the firstterminal of the capacitive device; and driving, with a controlled firstcurrent, the voltage of the second terminal of the precharged capacitivedevice in a direction to capacitively couple the first terminal and, byway of the current path, the common node to respective voltagessufficient in magnitude and polarity to cause a current, substantiallyequal to the first current, to flow collectively through one or bothtransistors of the differential transistor pair; wherein the firstcurrent flows as a displacement current through the capacitive deviceand is projected onto the common node, thereby providing a dynamic tailcurrent for the differential transistor pair substantially equal inmagnitude to the first current; developing a differential output signalat respective second current handling terminals of the differentialtransistor pair while the displacement current through the capacitivedevice provides a tail current for the differential transistor pair; andutilizing the differential output signal in a subsequent circuit coupledto the differential transistor pair; and wherein the differentialtransistor pair is incorporated within an I/O bus line amplifier of amemory array within the integrated circuit.
 27. In an integrated circuitincluding a differential transistor pair having a common node connectinga respective first current handling terminal of each transistor withinthe pair, a method of providing a tail current for the differentialtransistor pair comprising: precharging a capacitive device having afirst terminal and a second terminal; providing a current path from thecommon node of the differential transistor pair to the first terminal ofthe capacitive device; and driving, with a controlled first current, thevoltage of the second terminal of the precharged capacitive device in adirection to capacitively couple the first terminal and, by way of thecurrent path, the common node to respective voltages sufficient inmagnitude and polarity to cause a current, substantially equal to thefirst current, to flow collectively through one or both transistors ofthe differential transistor pair; wherein the first current flows as adisplacement current through the capacitive device and is projected ontothe common node, thereby providing a dynamic tail current for thedifferential transistor pair substantially equal in magnitude to thefirst current; developing a differential output signal at respectivesecond current handling terminals of the differential transistor pairwhile the displacement current through the capacitive device provides atail current for the differential transistor pair; and utilizing thedifferential output signal in a subsequent circuit coupled to thedifferential transistor pair; and wherein the differential transistorpair is incorporated within a bit line read amplifier of a memory arraywithin the integrated circuit.
 28. A method as in claim 1 wherein: thecapacitive device comprises a field effect transistor connected tofunction as a capacitor.
 29. A method as in claim 1 wherein: thecapacitive device comprises a capacitor.
 30. A method as in claim 1wherein: the capacitive device comprises a semiconductor junctionreversed biased to function as a capacitor.
 31. A method as in claim 1wherein the driving step comprises: driving the second terminal of thecapacitive device toward a power supply voltage.
 32. A method as inclaim 31 wherein the driving step further comprises: driving the secondterminal of the capacitive device through a switch transistor in serieswith a current source device.
 33. A method as in claim 32 wherein thecurrent source device comprises: a transistor biased with a referencevoltage coupled to the control terminal thereof.
 34. A method as inclaim 1 wherein: each transistor of the differential transistor paircomprises a field effect transistor and the common node comprises acommon-source node; while the first current flows through the capacitivedevice as a displacement current, the first terminal of the capacitivedevice assumes a voltage at which the current collectively flowingthrough one or both transistors of the differential transistor pairremains substantially equal to the first current; and for certain valuesof input voltages applied to the differential transistor pair, certaintransistor characteristics of the differential transistor pair, andcertain magnitudes of the first current, the voltage assumed by thefirst terminal of the capacitive device is outside a range of voltagesbounded by the highest power supply voltage and the lowest power supplyvoltage operably received by the integrated circuit.
 35. A method as inclaim 34 wherein: the capacitive device comprises a field effecttransistor connected to function as a capacitor and disposed in closelocal proximity to, and uniquely associated with, the differentialtransistor pair; and the current path includes at least one transistorconnecting the common node to the first terminal of the capacitivedevice, which at least one transistor is enabled to conduct currenttherebetween at least while driving the second terminal of thecapacitive device.
 36. In an integrated circuit including a differentialtransistor pair having a common node connecting a respective firstcurrent handling terminal of each transistor within the pair, a methodof providing a tail current for the differential transistor paircomprising: precharging a capacitive device having a first terminal anda second terminal; providing a current path from the common node of thedifferential transistor pair to the first terminal of the capacitivedevice; and driving, with a controlled first current, the voltage of thesecond terminal of the precharged capacitive device in a direction tocapacitively couple the first terminal and, by way of the current path,the common node to respective voltages sufficient in magnitude andpolarity to cause a current, substantially equal to the first current,to flow collectively through one or both transistors of the differentialtransistor pair; wherein the first current flows as a displacementcurrent through the capacitive device and is projected onto the commonnode, thereby providing a dynamic tail current for the differentialtransistor pair substantially equal in magnitude to the first current;wherein each transistor of the differential transistor pair comprises afield effect transistor and the common node comprises a common-sourcenode; wherein while the first current flows through the capacitivedevice as a displacement current, the first terminal of the capacitivedevice assumes a voltage at which the current collectively flowingthrough one or both transistors of the differential transistor pairremains substantially equal to the first current; and wherein forcertain values of input voltages applied to the differential transistorpair, certain transistor characteristics of the differential transistorpair, and certain magnitudes of the first current, the voltage assumedby the first terminal of the capacitive device is outside a range ofvoltages bounded by the highest power supply voltage and the lowestpower supply voltage operably received by the integrated circuit;wherein the capacitive device comprises a field effect transistorconnected to function as a capacitor and disposed in close localproximity to, and uniquely associated with, the differential transistorpair; wherein the current path includes at least one transistorconnecting the common node to the first terminal of the capacitivedevice, which at least one transistor is enabled to conduct currenttherebetween at least while driving the second terminal of thecapacitive device; said method further comprising: coupling an externalinput signal to a control terminal of one transistor of the differentialtransistor pair; and coupling a reference voltage to a control terminalof the other transistor of the differential transistor pair.
 37. Amethod as in claim 35 further comprising: coupling an external inputsignal to a control terminal of one transistor of the differentialtransistor pair; and coupling a complementary external input signal to acontrol terminal of the other transistor of the differential transistorpair.
 38. A method as in claim 34 wherein the driving step furthercomprises: driving the second terminal of the capacitive device througha switch transistor in series with a current source device.
 39. Anintegrated circuit comprising: a first transistor and a secondtransistor configured as a differential transistor pair having a commonnode connecting respective first current handling terminals of eachtransistor within the pair; capacitive means having a first terminal anda second terminal; means for precharging the capacitive means to avoltage thereacross; means for providing a current path from the commonnode of the differential transistor pair to the first terminal of thecapacitive means; and means for driving, with a substantially constantfirst current, the voltage of the second terminal of the prechargedcapacitive means in a direction to capacitively couple the firstterminal and, by way of the current path, the common node to respectivevoltages sufficient in magnitude and polarity to cause a current,substantially equal to the first current, to flow collectively throughone or both transistors of the differential transistor pair; wherein thefirst current flows as a displacement current through the capacitivemeans and is projected onto the common node, thereby providing a dynamictail current for the differential transistor pair substantially equal inmagnitude to the first current.
 40. An integrated circuit as in claim 39wherein: the voltage of the second terminal of the capacitive means,while the first current flows through the capacitive means as adisplacement current, changes at a rate that substantially exceeds therate of voltage change of the first terminal of the capacitive means.41. An integrated circuit as in claim 39 wherein: the capacitive meansis disposed in close proximity to, and uniquely associated with, thedifferential transistor pair.
 42. An integrated circuit as in claim 39wherein: the capacitive means comprises a field effect transistorconnected to function as a capacitor.
 43. An integrated circuit as inclaim 39 wherein: the capacitive means comprises a capacitor.
 44. Anintegrated circuit as in claim 39 wherein: the driving means comprises aswitch transistor in series with a current source device, togethercoupled between the second terminal of the capacitive means and a powersupply terminal.
 45. An integrated circuit comprising: a firsttransistor and a second transistor configured as a differentialtransistor pair having a common node connecting respective first currenthandling terminals of each transistor within the pair; capacitive meanshaving a first terminal and a second terminal; means for precharging thecapacitive means to a voltage thereacross; means for providing a currentpath from the common node of the differential transistor pair to thefirst terminal of the capacitive means; and means for driving, with afirst current, the voltage of the second terminal of the prechargedcapacitive means in a direction to capacitively couple the firstterminal and, by way of the current path, the common node to respectivevoltages sufficient in magnitude and polarity to cause a current,substantially equal to the first current, to flow collectively throughone or both transistors of the differential transistor pair; wherein thefirst current flows as a displacement current through the capacitivemeans and is projected onto the common node, thereby providing a dynamictail current for the differential transistor pair substantially equal inmagnitude to the first current; and wherein the current path providingmeans comprises a direct wired connection between the common node andthe first terminal of the capacitive means.
 46. An integrated circuit asin claim 39 wherein: the current path providing means comprises at leastone transistor coupled between the common node and the first terminal ofthe capacitive means, which transistor is conductive at least during atime when the driving means is enabled to drive the second terminal ofthe capacitive means.
 47. An integrated circuit as in claim 46 wherein:the at least one transistor coupled between the common node and thefirst terminal of the capacitive means is responsive to an enable signalwhich is active when the driving means is enabled.
 48. An integratedcircuit comprising: a first transistor and a second transistorconfigured as a differential transistor pair having a common nodeconnecting respective first current handling terminals of eachtransistor within the pair; a capacitive device having a first terminaland a second terminal; a precharge circuit for precharging, during afirst time period, the capacitive device to an initial voltagethereacross, said first terminal of the capacitive device being coupledto a source of a first voltage and said second terminal of thecapacitive device being coupled to a source of a second voltage; acurrent path from the common node of the differential transistor pair tothe first terminal of the capacitive device; and a driver circuitresponsive to a first control signal, for driving, with a substantiallyconstant first current during a second time period, the voltage of thesecond terminal of the capacitive device in a direction to capacitivelycouple the first terminal and, by way of the current path, the commonnode to respective voltages sufficient in magnitude and polarity tocause a current, substantially equal in magnitude to the first current,to flow collectively through one or both transistors of the differentialtransistor pair; wherein the first current flows as a displacementcurrent through the capacitive device and is projected onto the commonnode, thereby providing a dynamic tail current for the differentialtransistor pair substantially equal in magnitude to the first current.49. An integrated circuit as in claim 48 wherein: the voltage of thesecond terminal of the capacitive device, while the first current flowsthrough the capacitive device as a displacement current, changes at arate that substantially exceeds the rate of voltage change of the firstterminal of the capacitive device.
 50. An integrated circuit as in claim48 wherein: each transistor of the differential transistor paircomprises a field effect transistor; and the common node comprises acommon-source node.
 51. An integrated circuit as in claim 48 wherein:each transistor of the differential transistor pair comprises a bipolarjunction transistor; and the common node comprises a common-emitternode.
 52. An integrated circuit as in claim 48 wherein: each transistorof the differential transistor pair comprises a gallium arsenide MESFETtransistor; and the common node comprises a common-source node.
 53. Anintegrated circuit as in claim 48 wherein: while the first current flowsthrough the capacitive device as a displacement current, the firstterminal of the capacitive device assumes a voltage at which the currentcollectively flowing through one or both transistors of the differentialtransistor pair remains substantially equal to the first current.
 54. Anintegrated circuit comprising: a first transistor and a secondtransistor configured as a differential transistor pair having a commonnode connecting respective first current handling terminals of eachtransistor within the pair; a capacitive device having a first terminaland a second terminal; a precharge circuit for precharging, during afirst time period, the capacitive device to an initial voltagethereacross, said first terminal of the capacitive device being coupledto a source of a first voltage and said second terminal of thecapacitive device being coupled to a source of a second voltage; acurrent path from the common node of the differential transistor pair tothe first terminal of the capacitive device; and a driver circuitresponsive to a first control signal, for driving, with a first currentduring a second time period, the voltage of the second terminal of thecapacitive device in a direction to capacitively couple the firstterminal and, by way of the current path, the common node to respectivevoltages sufficient in magnitude and polarity to cause a current,substantially equal in magnitude to the first current, to flowcollectively through one or both transistors of the differentialtransistor pair; wherein the first current flows as a displacementcurrent through the capacitive device and is projected onto the commonnode, thereby providing a dynamic tail current for the differentialtransistor pair substantially equal in magnitude to the first current;wherein while the first current flows through the capacitive device as adisplacement current, the first terminal of the capacitive deviceassumes a voltage at which the current collectively flowing through oneor both transistors of the differential transistor pair remainssubstantially equal to the first current; and wherein the driver circuitis arranged to drive the second terminal of the capacitive device from avoltage substantially equal to a first power supply voltage operablyreceived by the integrated circuit toward a voltage substantially equalto a second power supply voltage operably received by the integratedcircuit.
 55. An integrated circuit comprising: a first transistor and asecond transistor configured as a differential transistor pair having acommon node connecting respective first current handling terminals ofeach transistor within the pair; a capacitive device having a firstterminal and a second terminal; a precharge circuit for precharging,during a first time period, the capacitive device to an initial voltagethereacross, said first terminal of the capacitive device being coupledto a source of a first voltage and said second terminal of thecapacitive device being coupled to a source of a second voltage; acurrent path from the common node of the differential transistor pair tothe first terminal of the capacitive device; and a driver circuitresponsive to a first control signal, for driving, with a first currentduring a second time period, the voltage of the second terminal of thecapacitive device in a direction to capacitively couple the firstterminal and, by way of the current path, the common node to respectivevoltages sufficient in magnitude and polarity to cause a current,substantially equal in magnitude to the first current, to flowcollectively through one or both transistors of the differentialtransistor pair; wherein the first current flows as a displacementcurrent through the capacitive device and is projected onto the commonnode, thereby providing a dynamic tail current for the differentialtransistor pair substantially equal in magnitude to the first current;wherein while the first current flows through the capacitive device as adisplacement current, the first terminal of the capacitive deviceassumes a voltage at which the current collectively flowing through oneor both transistors of the differential transistor pair remainssubstantially equal to the first current; and wherein the magnitude ofthe displacement current through the capacitive device is largelycontrolled by the output impedance of the driver circuit.
 56. Anintegrated circuit as in claim 48 wherein: the capacitive device isdisposed in close local proximity to, and uniquely associated with, thedifferential transistor pair.
 57. An integrated circuit comprising: afirst transistor and a second transistor configured as a differentialtransistor pair having a common node connecting respective first currenthandling terminals of each transistor within the pair; a capacitivedevice having a first terminal and a second terminal; a prechargecircuit for precharging, during a first time period, the capacitivedevice to an initial voltage thereacross, said first terminal of thecapacitive device being coupled to a source of a first voltage and saidsecond terminal of the capacitive device being coupled to a source of asecond voltage; a current path from the common node of the differentialtransistor pair to the first terminal of the capacitive device; and adriver circuit responsive to a first control signal, for driving, with afirst current during a second time period, the voltage of the secondterminal of the capacitive device in a direction to capacitively couplethe first terminal and, by way of the current path, the common node torespective voltages sufficient in magnitude and polarity to cause acurrent, substantially equal in magnitude to the first current, to flowcollectively through one or both transistors of the differentialtransistor pair; wherein the first current flows as a displacementcurrent through the capacitive device and is projected onto the commonnode, thereby providing a dynamic tail current for the differentialtransistor pair substantially equal in magnitude to the first current;said integrated circuit further comprising a second differentialtransistor pair; and wherein the capacitive device is disposed in closelocal proximity to, and also uniquely associated with, said differentialtransistor pair and said second differential transistor pair; and saiddifferential transistor pair and said second differential transistorpair each includes a respective gating circuit connecting the respectivecommon node thereof to the first terminal of the capacitive device. 58.An integrated circuit comprising: a first transistor and a secondtransistor configured as a differential transistor pair having a commonnode connecting respective first current handling terminals of eachtransistor within the pair; a capacitive device having a first terminaland a second terminal; a precharge circuit for precharging, during afirst time period, the capacitive device to an initial voltagethereacross, said first terminal of the capacitive device being coupledto a source of a first voltage and said second terminal of thecapacitive device being coupled to a source of a second voltage; acurrent path from the common node of the differential transistor pair tothe first terminal of the capacitive device; and a driver circuitresponsive to a first control signal, for driving, with a first currentduring a second time period, the voltage of the second terminal of thecapacitive device in a direction to capacitively couple the firstterminal and, by way of the current path, the common node to respectivevoltages sufficient in magnitude and polarity to cause a current,substantially equal in magnitude to the first current, to flowcollectively through one or both transistors of the differentialtransistor pair; wherein the first current flows as a displacementcurrent through the capacitive device and is projected onto the commonnode, thereby providing a dynamic tail current for the differentialtransistor pair substantially equal in magnitude to the first current;and wherein the current path consists of a direct wired connectionbetween the common node and the first terminal of the capacitive device.59. An integrated circuit comprising: a first transistor and a secondtransistor configured as a differential transistor pair having a commonnode connecting respective first current handling terminals of eachtransistor within the pair; a capacitive device having a first terminaland a second terminal; a precharge circuit for precharging, during afirst time period, the capacitive device to an initial voltagethereacross, said first terminal of the capacitive device being coupledto a source of a first voltage and said second terminal of thecapacitive device being coupled to a source of a second voltage; acurrent path from the common node of the differential transistor pair tothe first terminal of the capacitive device; and a driver circuitresponsive to a first control signal, for driving, with a first currentduring a second time period, the voltage of the second terminal of thecapacitive device in a direction to capacitively couple the firstterminal and, by way of the current path, the common node to respectivevoltages sufficient in magnitude and polarity to cause a current,substantially equal in magnitude to the first current, to flowcollectively through one or both transistors of the differentialtransistor pair; wherein the first current flows as a displacementcurrent through the capacitive device and is projected onto the commonnode, thereby providing a dynamic tail current for the differentialtransistor pair substantially equal in magnitude to the first current;and wherein the current path includes at least one transistor connectingthe common node to the first terminal of the capacitive device, whichcurrent path is enabled to conduct current therebetween at least duringa time when the driver circuit is enabled by the first control signal.60. An integrated circuit as in claim 59 wherein: the current pathincludes no device configured to function as a constant current sourceduring a time when the driver circuit is enabled by the first controlsignal; and the magnitude of the first current is controlled by thedriver circuit.
 61. An integrated circuit as in claim 59 wherein: thecurrent path includes at least one device configured to function as aconstant current source during a time when the driver circuit is enabledby the first control signal; and the magnitude of the first current iscontrolled by the constant current source.
 62. An integrated circuitcomprising: a first transistor and a second transistor configured as adifferential transistor pair having a common node connecting respectivefirst current handling terminals of each transistor within the pair; acapacitive device having a first terminal and a second terminal; aprecharge circuit for precharging, during a first time period, thecapacitive device to an initial voltage thereacross, said first terminalof the capacitive device being coupled to a source of a first voltageand said second terminal of the capacitive device being coupled to asource of a second voltage; a current path from the common node of thedifferential transistor pair to the first terminal of the capacitivedevice; and a driver circuit responsive to a first control signal, fordriving, with a first current during a second time period, the voltageof the second terminal of the capacitive device in a direction tocapacitively couple the first terminal and, by way of the current path,the common node to respective voltages sufficient in magnitude andpolarity to cause a current, substantially equal in magnitude to thefirst current, to flow collectively through one or both transistors ofthe differential transistor pair; wherein the first current flows as adisplacement current through the capacitive device and is projected ontothe common node, thereby providing a dynamic tail current for thedifferential transistor pair substantially equal in magnitude to thefirst current; and wherein the differential transistor pair isincorporated within a linear amplifier circuit.
 63. An integratedcircuit comprising: a first transistor and a second transistorconfigured as a differential transistor pair having a common nodeconnecting respective first current handling terminals of eachtransistor within the pair; a capacitive device having a first terminaland a second terminal; a precharge circuit for precharging, during afirst time period, the capacitive device to an initial voltagethereacross, said first terminal of the capacitive device being coupledto a source of a first voltage and said second terminal of thecapacitive device being coupled to a source of a second voltage; acurrent path from the common node of the differential transistor pair tothe first terminal of the capacitive device; and a driver circuitresponsive to a first control signal, for driving, with a first currentduring a second time period, the voltage of the second terminal of thecapacitive device in a direction to capacitively couple the firstterminal and, by way of the current path, the common node to respectivevoltages sufficient in magnitude and polarity to cause a current,substantially equal in magnitude to the first current, to flowcollectively through one or both transistors of the differentialtransistor pair; wherein the first current flows as a displacementcurrent through the capacitive device and is projected onto the commonnode, thereby providing a dynamic tail current for the differentialtransistor pair substantially equal in magnitude to the first current;and wherein the differential transistor pair is incorporated within alatching amplifier circuit having a separate latch enable input.
 64. Anintegrated circuit comprising: a first transistor and a secondtransistor configured as a differential transistor pair having a commonnode connecting respective first current handling terminals of eachtransistor within the pair; a capacitive device having a first terminaland a second terminal; a precharge circuit for precharging, during afirst time period, the capacitive device to an initial voltagethereacross, said first terminal of the capacitive device being coupledto a source of a first voltage and said second terminal of thecapacitive device being coupled to a source of a second voltage; acurrent path from the common node of the differential transistor pair tothe first terminal of the capacitive device; and a driver circuitresponsive to a first control signal, for driving, with a first currentduring a second time period, the voltage of the second terminal of thecapacitive device in a direction to capacitively couple the firstterminal and, by way of the current path, the common node to respectivevoltages sufficient in magnitude and polarity to cause a current,substantially equal in magnitude to the first current, to flowcollectively through one or both transistors of the differentialtransistor pair; wherein the first current flows as a displacementcurrent through the capacitive device and is projected onto the commonnode, thereby providing a dynamic tail current for the differentialtransistor pair substantially equal in magnitude to the first current;and wherein the differential transistor pair is incorporated within alatching amplifier circuit having no separate latch enable input; andwherein the latching amplifier circuit stores new information as aresult of the differential output signal developed by the differentialtransistor pair during the second time period.
 65. An integrated circuitas in claim 64 wherein the integrated circuit further comprises: a pairof cross-coupled inverters configured as a latch, each inverter's outputrespectively connected to the other inverter's input and also connectedto a respective second current handling terminal of the transistorswithin the differential transistor pair; wherein the first controlsignal provides a strobe signal for the latch.
 66. An integrated circuitas in claim 62 wherein the integrated circuit further comprises: a pairof load devices, each respectively connected between a respective secondcurrent handling terminal of the differential transistor pair and apower supply terminal.
 67. An integrated circuit comprising: a firsttransistor and a second transistor configured as a differentialtransistor pair having a common node connecting respective first currenthandling terminals of each transistor within the pair; a capacitivedevice having a first terminal and a second terminal; a prechargecircuit for precharging, during a first time period, the capacitivedevice to an initial voltage thereacross, said first terminal of thecapacitive device being coupled to a source of a first voltage and saidsecond terminal of the capacitive device being coupled to a source of asecond voltage; a current path from the common node of the differentialtransistor pair to the first terminal of the capacitive device; and adriver circuit responsive to a first control signal, for driving, with afirst current during a second time period, the voltage of the second.terminal of the capacitive device in a direction to capacitively couplethe first terminal and, by way of the current path, the common node torespective voltages sufficient in magnitude and polarity to cause acurrent, substantially equal in magnitude to the first current, to flowcollectively through one or both transistors of the differentialtransistor pair; wherein the first current flows as a displacementcurrent through the capacitive device and is projected onto the commonnode, thereby providing a dynamic tail current for the differentialtransistor pair substantially equal in magnitude to the first current;and wherein the differential transistor pair is incorporated within anexternal input buffer of the integrated circuit.
 68. An integratedcircuit as in claim 67 wherein: the first and second time periods arecontrolled to occur at respective times relative to an external clocksignal operably received by the integrated circuit.
 69. An integratedcircuit as in claim 67 wherein: a control terminal of one transistor ofthe differential transistor pair is coupled to receive an external inputsignal; and a control terminal of the other transistor of thedifferential transistor pair is coupled to receive a reference voltage.70. An integrated circuit as in claim 67 further comprising: a controlterminal of one transistor of the differential transistor pair iscoupled to receive an external input signal; and a control terminal ofthe other transistor of the differential transistor pair is coupled toreceive a complementary external input signal.
 71. An integrated circuitcomprising: a first transistor and a second transistor configured as adifferential transistor pair having a common node connecting respectivefirst current handling terminals of each transistor within the pair; acapacitive device having a first terminal and a second terminal; aprecharge circuit for precharging, during a first time period, thecapacitive device to an initial voltage thereacross, said first terminalof the capacitive device being coupled to a source of a first voltageand said second terminal of the capacitive device being coupled to asource of a second voltage; a current path from the common node of thedifferential transistor pair to the first terminal of the capacitivedevice; and a driver circuit responsive to a first control signal, fordriving, with a first current during a second time period, the voltageof the second terminal of the capacitive device in a direction tocapacitively couple the first terminal and, by way of the current path,the common node to respective voltages sufficient in magnitude andpolarity to cause a current, substantially equal in magnitude to thefirst current, to flow collectively through one or both transistors ofthe differential transistor pair; wherein the first current flows as adisplacement current through the capacitive device and is projected ontothe common node, thereby providing a dynamic tail current for thedifferential transistor pair substantially equal in magnitude to thefirst current; and wherein the differential transistor pair isincorporated within an I/O bus line amplifier of a memory array withinthe integrated circuit.
 72. An integrated circuit comprising: a firsttransistor and a second transistor configured as a differentialtransistor pair having a common node connecting respective first currenthandling terminals of each transistor within the pair; a capacitivedevice having a first terminal and a second terminal; a prechargecircuit for precharging, during a first time period, the capacitivedevice to an initial voltage thereacross, said first terminal of thecapacitive device being coupled to a source of a first voltage and saidsecond terminal of the capacitive device being coupled to a source of asecond voltage; a current path from the common node of the differentialtransistor pair to the first terminal of the capacitive device; and adriver circuit responsive to a first control signal, for driving, with afirst current during a second time period, the voltage of the secondterminal of the capacitive device in a direction to capacitively couplethe first terminal and, by way of the current path, the common node torespective voltages sufficient in magnitude and polarity to cause acurrent, substantially equal in magnitude to the first current, to flowcollectively through one or both transistors of the differentialtransistor pair; wherein the first current flows as a displacementcurrent through the capacitive device and is projected onto the commonnode, thereby providing a dynamic tail current for the differentialtransistor pair substantially equal in magnitude to the first current;and wherein the differential transistor pair is incorporated within abit line read amplifier of a memory array within the integrated circuit.73. An integrated circuit as in claim 48 wherein: the capacitive devicecomprises a field effect transistor connected to function as acapacitor.
 74. An integrated circuit as in claim 48 wherein: thecapacitive device comprises a capacitor.
 75. An integrated circuit as inclaim 48 wherein: the driver circuit is arranged to drive the secondterminal of the capacitive device toward a power supply voltage.
 76. Anintegrated circuit comprising: a first transistor and a secondtransistor configured as a differential transistor pair having a commonnode connecting respective first current handling terminals of eachtransistor within the pair; a capacitive device having a first terminaland a second terminal; a precharge circuit for precharging, during afirst time period, the capacitive device to an initial voltagethereacross, said first terminal of the capacitive device being coupledto a source of a first voltage and said second terminal of thecapacitive device being coupled to a source of a second voltage; acurrent path from the common node of the differential transistor pair tothe first terminal of the capacitive device; and a driver circuitresponsive to a first control signal, for driving, with a first currentduring a second time period, the voltage of the second terminal of thecapacitive device in a direction to capacitively couple the firstterminal and, by way of the current path, the common node to respectivevoltages sufficient in magnitude and polarity to cause a current,substantially equal in magnitude to the first current, to flowcollectively through one or both transistors of the differentialtransistor pair; wherein the first current flows as a displacementcurrent through the capacitive device and is projected onto the commonnode, thereby providing a dynamic tail current for the differentialtransistor pair substantially equal in magnitude to the first current;wherein the driver circuit is arranged to drive the second terminal ofthe capacitive device toward a power supply voltage; and wherein thedriver circuit is arranged to drive the second terminal of thecapacitive device with a constant current.
 77. An integrated circuit asin claim 76 wherein the driver circuit comprises: a switch transistor inseries with a current source device, together coupled between the secondterminal of the capacitive device and a power supply terminal operablycoupled to receive the power supply voltage.
 78. An integrated circuitas in claim 77 wherein the current source device comprises: a transistorbiased with a reference voltage coupled to the control terminal thereof.79. An integrated circuit as in claim 48 wherein: each transistor of thedifferential transistor pair comprises a field effect transistor and thecommon node comprises a common-source node; and while the first currentflows through the capacitive device as a displacement current, the firstterminal of the capacitive device assumes a voltage at which the currentcollectively flowing through one or both transistors of the differentialtransistor pair remains substantially equal to the first current.
 80. Anintegrated circuit comprising: a first transistor and a secondtransistor configured as a differential transistor pair having a commonnode connecting respective first current handling terminals of eachtransistor within the pair; a capacitive device having a first terminaland a second terminal; a precharge circuit for precharging, during afirst time period, the capacitive device to an initial voltagethereacross, said first terminal of the capacitive device being coupledto a source of a first voltage and said second terminal of thecapacitive device being coupled to a source of a second voltage; acurrent path from the common node of the differential transistor pair tothe first terminal of the capacitive device; and a driver circuitresponsive to a first control signal, for driving, with a first currentduring a second time period, the voltage of the second terminal of thecapacitive device in a direction to capacitively couple the firstterminal and, by way of the current path, the common node to respectivevoltages sufficient in magnitude and polarity to cause a current,substantially equal in magnitude to the first current, to flowcollectively through one or both transistors of the differentialtransistor pair; wherein the first current flows as a displacementcurrent through the capacitive device and is projected onto the commonnode, thereby providing a dynamic tail current for the differentialtransistor pair substantially equal in magnitude to the first current;wherein each transistor of the differential transistor pair comprises afield effect transistor and the common node comprises a common-sourcenode; wherein while the first current flows through the capacitivedevice as a displacement current, the first terminal of the capacitivedevice assumes a voltage at which the current collectively flowingthrough one or both transistors of the differential transistor pairremains substantially equal to the first current; wherein the capacitivedevice comprises a field effect transistor connected to function as acapacitor and disposed in close local proximity to, and uniquelyassociated with, the differential transistor pair; and wherein thecurrent path includes at least one transistor connecting the common nodeto the first terminal of the capacitive device, which transistor isenabled to conduct current therebetween at least during a time when thedriver circuit is enabled by the first control signal.
 81. An integratedcircuit as in claim 80 wherein: the differential transistor pair isincorporated within an external input buffer of the integrated circuit;a control terminal of one transistor of the differential transistor pairis coupled to receive an external input signal; and a control terminalof the other transistor of the differential transistor pair is coupledto receive a reference voltage.
 82. An integrated circuit as in claim 80wherein: the differential transistor pair is incorporated within anexternal input buffer of the integrated circuit; a control terminal ofone transistor of the differential transistor pair is coupled to receivean external input signal; and a control terminal of the other transistorof the differential transistor pair is coupled to receive acomplementary external input signal.
 83. An integrated circuit as inclaim 80 wherein the driver circuit comprises: a switch transistor inseries with a current source device, together coupled between the secondterminal of the capacitive device and a power supply terminal.
 84. In anintegrated circuit including a first NMOS transistor and a second NMOStransistor configured as a differential transistor pair and havingcommonly-connected source terminals, said differential transistor pairfor developing a differential current in response to a differentialvoltage applied between the respective gate terminals of the first andsecond NMOS transistors, a method of providing a tail current throughthe differential transistor pair, said method comprising: precharging acapacitive device by coupling a first terminal thereof to a firstvoltage and coupling a second terminal thereof to a second voltagehigher than the first voltage; then driving, with a substantiallyconstant first current, the second terminal of the precharged capacitivedevice toward a lower voltage, thereby tending to capacitively couplethe first terminal of the capacitive device toward a voltage below thefirst voltage; and providing a current path from the common source nodeto the first terminal of the capacitive device at least during a timewhen the first terminal if capacitively coupled toward a lower voltage;wherein the first terminal and, by way of the current path, the commonnode assume respective voltages sufficiently low to cause a current,substantially equal to the first current, to flow collectively throughone or both transistors of the differential transistor pair; and whereinthe first current flows as a displacement current through the capacitivedevice and is projected onto the common node, thereby providing adynamic tail current for the differential transistor pair substantiallyequal in magnitude to the first current.
 85. A method as in claim 84wherein: for certain values of input voltages applied to thedifferential transistor pair, certain transistor characteristics of thedifferential transistor pair, and certain magnitudes of the firstcurrent, the voltage assumed by the common source node is below thelowest power supply voltage operably received by the integrated circuit.86. In an integrated circuit including a first NMOS transistor and asecond NMOS transistor configured as a differential transistor pair andhaving commonly-connected source terminals, said differential transistorpair for developing a differential current in response to a differentialvoltage applied between the respective gate terminals of the first andsecond NMOS transistors, a method of providing a tail current throughthe differential transistor pair, said method comprising: precharging acapacitive device by coupling a first terminal thereof to a firstvoltage and coupling a second terminal thereof to a second voltagehigher than the first voltage; then driving, with a controlled firstcurrent, the second terminal of the precharged capacitive device towarda lower voltage, thereby tending to capacitively couple the firstterminal of the capacitive device toward a voltage below the firstvoltage; and providing a current path from the common source node to thefirst terminal of the capacitive device at least during a time when thefirst terminal is capacitively coupled toward a lower voltage; whereinthe first terminal and, by way of the current path, the common nodeassume respective voltages sufficiently low to cause a current,substantially equal to the first current, to flow collectively throughone or both transistors of the differential transistor pair; wherein thefirst current flows as a displacement current through the capacitivedevice and is projected onto the common node, thereby providing adynamic tail current for the differential transistor pair substantiallyequal in magnitude to the first current; and wherein for certain valuesof input voltages applied to the differential transistor pair, certaintransistor characteristics of the differential transistor pair, andcertain magnitudes of the first current, the voltage assumed by thecommon source node is above the lowest power supply voltage operablyreceived by the integrated circuit.